Author Topic: Adventures in induction heating  (Read 1704 times)

Offline Anders Mikkelsen

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Adventures in induction heating
« on: August 22, 2017, 11:30:04 PM »
Hello All,

This thread will be used to document a project that has been ongoing for a few years now, on and off, and part of it has already been documented on 4hv. I will start by reposting the 4hv material, then I will try to document the one and a half years of progress since I stopped updating the thread.

To give a quick introduction, the idea was to build a workshop induction heater, capable of pulling as much power as is available from a 230 V 16 A mains outlet. It should be robust enough to survive being used by somebody who didn't pay for it, and simple enough to not warrant a manual. All parts used in the design should be in current production and not unreasonably expensive. Output power should be adjustable, and at the maximum setting, it should be able to deliver at least half of its rated power into its own work-coil (given a typical solenoidal work coil of reasonable design). The frequency range of interest is 100 - 500 kHz, to be able to use commonly available surplus conduction cooled capacitors. An ambitious set of goals, but how else to keep it interesting?

So where am I now? The latest prototype fulfills all the goals, aside from the power delivery, which hasn't been fully tested. I also spent months researching and testing tank circuits, matching networks and transformers, this will be documented in this thread in the future. Other projects and work took over priority at that point, but after having spent almost a year developing SiC-based high power converters, I have some new ideas and inspiration to continue the project, with slightly revised goals; mainly increased power and 3 phase 400 V support, which basically comes for free when using 900 V SiC devices.

To not leave this first post as a wall of text, I'll include a picture of some random melting experiments from an early stage of the project

Offline Anders Mikkelsen

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Re: Adventures in induction heating
« Reply #1 on: August 24, 2017, 07:48:25 PM »
January 2015:

This thread is both to document my experiences and experiments in induction heating, and to present what I have learned along the way. Like so many hobby projects, this one has a very real chance of ending up as an unfinished pile of components, so by documenting what I've done so far, at least some use can come out of it. Along the way, a lot of my preconceptions about induction heating have been proven wrong, so by sharing what I've learned, I can hopefully dispel some common myths. My ultimate goal is to create a compact stand-alone induction heater circuit that can drive a variety of tank circuits, both LCLR and series resonant, at up to 500 kHz and possibly higher. It should be able to handle as much power as a regular european outlet can supply, that is around 20A at 230V, and the power level should be adjustable, with a power factor close to unity. Further, it should be robust enough to be used in a workshop setting for normal everyday tasks. Lastly, it should not use any exotic or expensive components, aside from the tank capacitor. From my research so far, I believe this is possible, but not trivial.

To not lose everybody's interest by starting with theory, I'll begin with the results of my experiments so far. To get a feeling of how everything works together, I made a simple open-loop LCLR half bridge induction heater. The tank circuit was construced from a Celem CP80/200 400nF 500V 400A mica capacitor and a 500nH workcoil made from 5mm copper pipe. An 8µH matching inductor made from CAT5 network cable connected the tank circuit to a MOSFET half bridge. The MOSFETs were driven by a pair of IXDN614 gate drivers, through a gate drive transformer. An open loop oscillator made from the VCO section of a 74HC4046 provided the operating frequency. Power for the half-bridge was provided by an isolated 3A Variac, and a bridge rectifier. DC bus capacitors were kept minimal to ensure a good power factor. This has an added advantage of keeping the stored energy low, minimizing any damage in case of a MOSFET failure. Here is a video giving an overview of the test setup


So far, the circuit has performed well, and it has given me a good feel of how the induction heater responds to different load conditions and operating frequencies. The tank circuit resonates at around 360 kHz unloaded, and I've had it up to 400kHz when loaded with large conductive objects. I've mostly limited myself to the 3A that the variac is rated for, .but during some more enthusiastic moments I've had the 3A meter needle pegged, probably at somewhere north of 4A (at 250V AC). It melts thin pieces of steel without any problem, here is a video of it melting a bracket from a network card, which is made of 1mm thick steel


The MOSFETs I used had about 10nF of gate capacitance each, and at the frequencies I'm using it's difficult to get reasonable rise times with a GDT. Therefore, it has been difficult to introduce dead time to get ZVS. The MOSFETs also have significant conduction losses, even though they are some of the lowest Rdson 500V MOSFETs I could find. IGBTs would fix both these problems, so for the next prototype I'm going for IGBTs. The downside with IGBTs is tail current losses, which can be fixed by ensuring ZCS. The easiest way to do this is by switching them at the tank resonant frequency. This rules out power control by detuning, further complicating things.

The next post will be about my prototype mark II, or maybe some theory.
« Last Edit: August 24, 2017, 08:03:49 PM by Anders Mikkelsen »

Offline Anders Mikkelsen

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Re: Adventures in induction heating
« Reply #2 on: August 24, 2017, 07:51:05 PM »
February 2015:

For Mk II of the prototype, I'm going with a transformer coupled series resonant tank circuit. Not neccessarily because I think it's superior to LCLR in any way, but I want to try it as a fair comparison.

The driver is completely revised. A 4046 PLL is used to control the switching, with a loop filter design stolen from Steve Conner's DRSSTC controller. Phase comparator 1 is used, due to its good immunity to noise. The PLL locks the phase angle between the VCO and the power capacitor voltage, with a selectable phase offset. This allows the switching phase angle to be adjusted between early and late, allowing both ZVS and ZCS.



The MOSFET halfbridge has now been replaced with an IGBT fullbridge using some new cheap Fairchild FGT40T65SPDs on a common heatsink, isolated with kapton tape. The heatsink is an old Intel Pentium 4 one with a small fan, it's a bit on the small side, but its small thermal mass makes it easy to get an idea of the IGBT power dissipation as a function of switching phase angle and loading. As mentioned in the previous post, the IGBTs have much lower input capacitance, 1.4 nF vs 10 nF, and conduction losses are significantly lower. IGBTs generally have much higher switching loss, but the use of ZVS and/or ZCS should help combat this. Whether this works well in practice remains to be seen, but preliminary tests look very promising.

The IGBTs are driven by Silabs SI8234 isolated 4A drivers. These drivers are extremely practical, with internal dead-time generation and UVLO on both gate drive outputs. As opposed to the classic high-side drivers like the IR2110, both drive outputs are isolated, and completely isolated from the input side. This allows the control electronics to be completely isolated from the mains side stuff, and unlike the classic high side drivers, it is completely immune to damage from the bridge midpoint swinging below ground, and bridge failures are not likely to kill the low voltage control electronics. Power to the upper driver is provided through a single bootstrap diode.

A classic problem with voltage fed series resonant IHs is that the power draw is maximum when unloaded, and lower when heavily loaded. Some designs are made to tolerate this condition. The disadvantage to this is that power transfer to a load will always be lower than the idle draw, and the tank VARs are badly utilized. Some designs use detuning to keep power in control. This works very well, but ZCS is lost, which can be a disadvantage when using IGBTs, especially at higher operating frequencies.

Power control is achieved through pulse skipping. As opposed to detuning, this allows ZVS to be maintained while reducing power. A second advantage to this method is that the effective switching frequency is reduced when it's skipping pulses, further lowering switching losses. The pulse skipping is implemented by sensing the tank voltage with a comparator, which triggers a flip flop that ensures that only whole cycles are skipped. As there is a 90 degree phase shift between the bridge output and the capacitor voltage, the flip flop will sample the comparator output around the peak of the tank circuit voltage. The idea of using pulse skipping is taken from an excellent paper I found  . Of course this is what Steve Conner has been doing all along in his DRSSTC driver. Just like many times before, I could have saved myself a lot of work if I used his excellent design in the first place.

One potential problem with this power control scheme is that it reacts very fast. This is generally a good thing, but when running from unsmoothed rectified mains it leads to a big problem. Since the loop is so fast, it will try to maintain a set voltage on the capacitor, so it will draw more current as the mains voltage drops. This will lead to horrible distortion of the mains current waveform and correspondingly a very bad power factor. The common solution is to slow down the feedback loop so that it rejects the mains frequency. Since the pulse skipping method needs to be fast to work properly, this is not an option here. There simplest solution to this problem is to take the setpoint for the pulse skipping comparator as a fraction of the DC bus voltage. This way, the current draw from mains will track the voltage, ensuring unity power factor.

One of the main questions I wanted to answer with this prototype was if it is possible to maintain ZVS and ZCS under all loading conditions with a simple PLL. So far it looks like the answer is no. When the phase angle is tuned for ZVS at light loading, ZVS is lost at heavy loading. If the switching phase angle is tuned for ZCS, it keeps switching at zero current independent of the loading. The ultimate goal is to minimize IGBT losses, so the best solution is anyways to tune the switching angle for lowest losses under normal operating conditions. Preliminary tests comparing ZVS and ZCS at low power (400 W input) show pretty similar IGBT losses, I will update the thread with results as I go higher in power.

Here's a schematic of the prototype. The protection circuitry is not implemented yet, and the comparator setpoint comes from the +5V rather than the DC bus.

« Last Edit: August 24, 2017, 08:04:05 PM by Anders Mikkelsen »

Offline Anders Mikkelsen

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Re: Adventures in induction heating
« Reply #3 on: August 24, 2017, 07:52:49 PM »
February 2015:

I've implemented pulse skipping for power control now. Both the lower IGBTs in the bridge are turned on for a whole cycle as long as the tank capacitor voltage exceeds the setpoint. Otherwise, the bridge outputs are driven by a 4046 PLL to maintain ZCS.

I've had loads of trouble with stability. I ran the induction heater from unsmoothed rectified mains, and set the pulse skip threshold as a DC value to cut into the peaks of the mains waveform. When the tank voltage exceeds the threshold, it starts pulse skipping but goes into some awful oscillations. The following picture shows the bridge outputs during this oscillation, at 20 microseconds per division. The DC bus voltage oscillates wildly, and actually tries to go below zero. Sorry about the crappy picture, I didn't have a USB stick handy to do a proper oscilloscope screenshot.



When running from smoothed rectified mains, the pulse skipping circuit works perfectly, and controls power smoothly from zero to full power. It also regulates instantly with load changes, as it should, and runs nicely and quietly. Here's a video showing both fullbridge outputs along with the inverter current (which is just a scaled version of the tank current due to the transformer fed series resonant tank circuit). In the beginning of the video, the tank is unloaded. A graphite crucible is put into the workcoil and removed again to show the effect of loading.


My understanding of control theory falls apart when I try to analyze where the problem is, as I'm having a hard time understanding where my phase margin exists in the circuit. I'm guessing that it's related to the time constant formed by the output impedance of my Variac and the DC bus cap, as the problem goes away when I add a lot of DC bus capacitance. Is there any way to make it more stable without adding said DC bus capacitors, or is there anything else I can do?

I considered the energy transfer between the tank and the bus, but I think this won't be a problem, as the bridge shorts out the load when skipping pulses.

I think mains inductance has something to do with the problem, but I can't really wrap my head around how it ends up oscillating. The problem could be the apparent negative input resistance of the bridge interacting with the bus capacitance and mains inductance. The circuit tries to maintain a constant voltage in the tank circuit, so the input power is constant, and input current sinks with rising input voltage.

In this case, the solution should be simple. Making the pulse skipping threshold track the mains voltage will invert the apparent bridge input impedance back to positive and stabilize it.

Noise doesn't seem to be a big problem so far.

I considered phase angle control with SCRs, but this will ruin the mains power factor.
« Last Edit: August 24, 2017, 08:04:27 PM by Anders Mikkelsen »

Offline Anders Mikkelsen

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Re: Adventures in induction heating
« Reply #4 on: August 24, 2017, 07:54:08 PM »
February 2015:

I did end up figuring out the problem with stability, and confirmed it by fixing the problem. It's now stable under all load conditions, and I still haven't managed to break a single IGBT, or any other component for that matter.

With tank voltage control by pulse skipping, the input resistance of the bridge looks negative. As the control circuit tries to maintain a constant voltage across the tank, the power input to the bridge is constant. This means that as the bridge input voltage rises, the current draw goes down, essentially making it look like a negative resistance. This negative resistance can excite resonances in the circuit, which is exactly what happened in my case.

At first, I though the rectifier bridge would eliminate potential resonances between the DC bus capacitors and the mains inductance, like the de-Qing diode in a DC charging Tesla coil, but this is not the case. Mains keeps the diodes forward biased, and allows the DC bus cap to swing with mains inductance. I got the first clue towards this when I tried resonating my DC bus cap with the mains input. The oscillation I saw matched the frequency of the instability. I discussed this with Steve Conner, and he agreed with my interpretation of the problem, and gave me a lot more insight in the process.

To solve this problem is easy, we only need to make the bridge input appear like a positive resistance to dampen any resonance here. Making the tank capacitor voltage track the bridge DC bus voltage will accomplish this nicely, and we need to do this anyways to get reasonable mains power factor. I already tried this earlier, without any success, but I didn't understand the problem properly and made a big mistake. Bridge voltage feedback was taken through a mains transformer for isolation, and the transformer has a low pass characteristic formed by the leakage inductance, stray capacitance and load resistance. This low pass characteristic meant that it only made the bridge input resistance look positive at low frequencies, but not at the resonance frequency of the bus caps and the mains inductance.

Once I understood the problem, I tried again, but without any bandwidth limiting components in the bridge voltage sensing loop. This worked perfectly, and the waveforms are very clean and everything is stable now. I have smooth power control from almost zero up to very high powers, and IGBT losses are as low as they can be. Mains power factor is very good, and the tank current tracks mains current very cleanly without much ripples. I have full control of the switching phase angle through the PLL, and I can tune it for ZVS, ZCS and potentially both. Below is a video of it in operation, with the scope showing the operational waveforms when tuned to ZVS and close to ZCS.


Right now the prototype is limited by heating in the coupling transformer primary. I use two parallel strands of ~1mm dia. teflon insulated silver plated wire, and it gets hot enough to boil water in a few minutes even at moderate power input. The next step is to make a proper PCB for it, and implement inverter overcurrent shutdown to protect against workcoil shorts. Next will be some experiments with using RF powder cores as matching inductors for LCLR. I also have some theory written up that I'll post in this thread when time allows.
« Last Edit: August 24, 2017, 08:04:52 PM by Anders Mikkelsen »

Offline Anders Mikkelsen

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Re: Adventures in induction heating
« Reply #5 on: August 24, 2017, 08:03:09 PM »
March 2015:

Some news on the project. I tried using the induction heater for practical stuff. The biggest limitation now is power dissipation in the coupling transformer primary. When running for a few minutes earlier, it got quite hot, and when trying to run it for longer this week, the insulation on the winding melted and the primary shorted, taking out the IGBTs. The wire I used was teflon insulated stranded 16 AWG silver plated, with two wires in parallel, and when it failed it was carrying around 10 A RMS at 300 kHz.



I suspected the skin effect losses were significant, so I tried replacing it with some heavy litz from an induction stove, 144 strands of 0.25mm for a total cross section of 7 mm^2. This is not ideal for the frequency in question, but should be significantly better than the smaller non-litz wire I used before in terms of skin effect losses. When testing it, there was hardly any improvement. It turns out most of the copper losses were caused by the proximity effect, especially near the secondary where the current density is very high. From a paper I found: "Simply increasing the wire size won't help; unlike skin effect, a larger than optimum wire size can dramatically increase the losses, especially for multiple winding layers. Litz wire is not a panacea and may also increase losses" http://energylogix.ca/losses_discussion.pdf



To try to minimize this problem, I soldered some copper plates to the secondary. These should hopefully spread out the proximity losses, to avoid overheating the primary near the secondary, and also help with cooling the primary by heat conduction to the water cooled secondary. If needed, the primary can be potted in thermally conductive glue, to help heat conduction to the secondary. I haven't tested it yet, but will update the thread with results


Offline Anders Mikkelsen

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Re: Adventures in induction heating
« Reply #6 on: August 24, 2017, 08:05:52 PM »
April 2015:

The transformer design is one of the most challenging parts. I think a successful design needs to both minimize dissipation and get rid of the heat that is produced in the windings and the core. Understanding the proximity effect is the dominating challenge for me now, I know my new transformer is a lot better but I don't know if it's good enough yet. If it isn't, then thermal glue and foil winding is the next step.

Non-isolated LCLR avoids this problem, and also has some other advantages. I've already written up something about the relative advantages and disadvantages of the different tank circuit arrangements:

A lot has been said about the relative merits of TCSR (transformer coupled series resonant) and LCLR tank circuits. I've seen statements that TCSR has a better power factor or better efficiency than LCLR. This is however not true, and both configurations are pretty similar in practice. I've melted steel with both, and there was no real difference in performance. This is assuming that the matching network is chosen properly, and I assume that people who have had bad experiences with LCLR have had badly chosen component values in the matching network. Theory and simulations also back this up, the phase and amplitude response is very similar between the two configurations near the resonant frequency. Both have a phase angle that crosses through zero at the resonant frequency, so both will have unity power factor at some frequency. Both are also used industrially. There are hower some practical differences between them, which I will try to sum up here.

Edit March 2016: I've done some more reading and simulations, and I have to add some corrections to this. What I wrote about the power factor of LCLR tank circuits only applies at light loading. At heavier loading of the work coil, the inverter current phase angle never crosses zero. This means that the LCLR has worse power factor and no possibility of ZCS with heavier loading.

LCLR advantages:
*Tolerant of work coil shorts, inverter sees a light inductive load
*Can be made without ferrite in the power circuit
*Little added parasitic inductance between the tank capacitor and the work coil, this can be very important with low inductance work coils
*Wiring inductance between inverter and tank circuit will just add to the matching inductor
*Easy to drive a tank from multiple bridges combined through matching inductors
*If the matching inductor is placed near the bridge, the transmission line voltage is sinusoidal, resulting in less radiated noise when the tank is far away
*Mechanically nice tank arrangement

LCLR disadvantages:
*Bad power factor with heavy loading
*No galvanic isolation without an extra transformer
*Requires a matching inductor that can also induction heat surrounding metal
*Current phase angle is not monotonic below resonance, making it hard for a simple PLL to always find resonance
*Matching steals a small part of the tank capacitance. Not usually a big problem

TCSR advantages:
*Unity power factor is possible over a wide range of loading contitions
*Inherent galvanic isolation
*No matching inductor is required
*The phase angle of the inverter current is always monotonic on both sides of resonance, so inverter current feedback can be used

TCSR disadvantages:
*Can be mechanically awkward to make the tank circuit
*Inverter is presented with a capacitive load if workcoil is shorted
*Inductance between the inverter and tank is reflected through the matching transformer, stealing tank VARs.
*Isolation transformer design is difficult at higher frequencies and powers, mostly due to the proximity effect, transformer coupled LCLR has similar challenges.

----------------------------------------------

The power control method is pretty simple. In my implementation, I use the sensed voltage from the tank cap to control the skipping. Whenever the capacitor voltage is higher than the reference, the bridge outputs are disabled. The number of cycles that will be skipped depends on the reference and how much load is placed in the heater. You need to make sure that it skips an even number of cycles so that you have the same number of positive and negative cycles. If you clock the JK flip flop from the same signal that controls the bridge, it will always skip an even number of cycles.

You also need to make sure that the phasing between the sense signal (tank voltage or current) is matched to when you need the signal. The tank voltage is shifted 90 degrees in relation to the bridge output voltage. The JK flip flop samples on the rising edge of one of the bridge outputs, so exactly when the cap voltage peaks. This means you can connect your comparator directly to the JK flip flop. I used this method originally, but there is some risk of metastability if the flip flop is clocked exactly when the comparator output changes state, so I went to a different method.

If you use tank or inverter current for feedback, the bridge output switching matches with the inverter/tank current zero crossing, so you need a second flip flop to hold the comparator state until the "Enable" flip flop can sample it. This method can also be used with tank voltage sensing, and it will solve the metastability problem. This is what I'm doing now, and you can see it in my latest schematic.

If you have little smoothing on the DC bus, it is important that you get the comparator reference voltage from the DC bus voltage. This will ensure that the circuit will be stable, and it will also ensure good power factor. If you have large electrolytics across the DC bus, it is not that critical, but still a very good idea.

For skipping pulses, I turn on both bottom switches to short out the tank circuit. This traps energy in the tank. This is not possible to do with a half-bridge, but you can open both switches to recycle the energy back to the DC bus, and it will still work fine. This is what Steve Conner does in his PLL DRSSTC driver.

Offline Anders Mikkelsen

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Re: Adventures in induction heating
« Reply #7 on: August 24, 2017, 08:09:27 PM »
May 2015:

There have been some updates on the project. The prototype was getting a bit complex, so I designed and ordered a board for it. With the newest changes, it's a bit closer to Steve Conner's DRSSTC driver. Now I take PLL feedback from the tank current intstead of the voltage. As the current should be in phase with the bridge voltage, I needed to cancel out the 90 degree shift from the XOR phase comparator. I used Steve's trick here, where I make the PLL run at 2x Fres by placing a divider in the feedback loop, and use a divider with an inverted clock to get back to Fres with an added 90 degree phase shift. This adds up to a 180 degree phase shift, which can easily be cancelled by inverting the signal.



The boards I designed 2 layers, and they include the latest feedback, bus voltage sensing and additional protection circuits. If everything works as expected, I already have a 4 layer design with more improvements ready to go to production. The 4-layer design should provide much better current handling for the bridge, and better signal integrity as well. The board is designed to have all the essential functions on-board, including the power supply. The only thing needed to make them run is to connect a potentiometer for power control and a few buttons and LEDs for start/stop and status. The control port also includes other signals like tank and mains current sensing, mains voltage sensing, VCO frequency and others, so that a more elaborate control panel can be made at a later point. All protection functions are inherent, so it will run safely with the minimal control interface.



As for the coupling transformer, I did some reading and simulations on the proximity effect, and it turns out that using conductors thicker than a fraction of the skin depth will radically increase the effective AC resistance, by as much as a factor of 100. Skin effect tells us that there's little benefit from any conductor over a given size (aside from increased surface area and thermal mass), but I didn't expect a too large conductor to make things that much worse. I did some simulations in Matlab, the following plot shows the effective AC resistance of a 20 layer foil primary as a function of the foil thickness. The green line is the DC resistance and the blue line is the AC ressitance at 300kHz. Notice how it goes up drastically beyond the ideal thickness. Luckily, the ideal thickness is the same as the thickness of regular copper tape.


Offline Anders Mikkelsen

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Re: Adventures in induction heating
« Reply #8 on: August 24, 2017, 08:10:20 PM »
June 2015:

It's been a while since I updated this thread. The project hasn't been inactive though, it's just hard to find time to sit down and document it these days.

A lot has happened in the meantime. I received the prototype boards a few weeks after ordering them, and managed to put one together. Some bodges with magnet wire were needed, as the boards were submitted to production before mister_rf found my flip-flop wiring bug. Everything works correctly, including the pulse density modulation and overcurrent detection, but there are still some things I want to improve. There are mainly two issues which i'll describe here.

The first issue is that the PLL easily loses lock. Especially at low bus voltage, low pulse density settings and when the PLL is set to switch early. This was never a problem in my first prototype that used capacitor voltage feedback, so it must be fixable. I did some modifications that improved the situation, but it still loses lock under some conditions so I'm working through the circuit step by step to make it more stable. The main modification that helped was to limit the upper and lower frequency of the PLL. The lower frequency can easily be changed with the resistor connected to pin 12 of the 4046, and the upper frequency can be limited by a voltage divider between the loop filter and the VCO input. Limiting the minimum power setting also helps. I'm slowly working through the circuit, using the totally stable prototype 1 as a comparison.

The second problem is more subtle and more fundamental. It's not really a big problem, so I didn't notice it until now, but I still want to fix it. The symptom is that the switching phase angle changes with the pulse density setting. The change is not huge, and the target phase can be set so that soft switching is maintained down to some reasonably low pulse density setting. The reason for the problem was a bit more mysterious, but again Steve Conner gave me some ideas. The fundamental problem is that the PLL loop dynamics change when pulses are skipped. During this idle time, the PLL is sensing tank current but not controlling it. For my circuit to work properly, the PLL loop needs to be stable both when the loop is open and closed. I've only designed the loop filter for stability when running closed loop. Indeed, the integrator loop filter will be unstable when running open loop, and the frequency drifts during the idle time. Now I need to figure out a loop filter to make it stable in both conditions. Most likely, this will only need an extra resistor and maybe a cap, but I haven't dug that deep into the theory yet. Hopefully, fixing this problem will also help with the first problem.

The values that need tweaking in my schematic are the following: The low pass filter for the bus voltage sensing needs to suppress switching noise but still be fast enough to stabilise the bus voltage. I ended up using 10k/100p , but even more capacitance could help when using lower switching frequencies. The dead-time should be set for a few hundred nanoseconds as a starting point. Target phase needs to be tweaked until it locks and then for ZVS and/or ZCS.

Offline Anders Mikkelsen

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Re: Adventures in induction heating
« Reply #9 on: August 24, 2017, 08:15:55 PM »
July 2015:

I've now figured out both problems from the previous post.

Part of the lock problem was caused by strange behavior of the 74HC4046 VCO outside of the recommended input voltage range. The TI datasheet recommends staying between 1 and 4 V for VCOin. My circuit does not have a restriction on the VCO input voltage, so it can potentially be between 0 and 5 V. The chip I used for the first prototype behaved nicely from 0 to 5 V, but for the second prototype I used a 74HC4046 from TI, which does some very strange things. Below 0.8 V in, the frequency steps abruptly to half of the value, and any further lowering of the control voltage has no effect at all. Increasing the control voltage above 4 V leads to an exponential rise in the output frequency until it hits 10 + MHz at 5V (with a 1 nF timing cap). Needless to say, this does some strange things to the loop dynamics, and the thing was hard to get stable, especially when the resonance frequency ends up in the middle of the step around 0.8 V. I added a resistor network, with 12k to VCC, 12k to GND and 6k8 to the loop filter output, all connected to VCOin. This made it behave a lot better.

I was still not entirely happy with the PLL current feedback. It's hard to get nice feedback with low phase shift when far from the resonant frequency. I experimented a bit more with capacitor voltage feedback, and it's a lot easier to get working well. Since I want to use capacitor voltage sensing for capacitor protection, there's no disadvantage to changing back to capacitor voltage feedback. After playing a bit with LTSpice, I managed to make a nice network for measuring average cap voltage and phase, without introducing unreasonable phase shifts or zero crossing asymmetry over a wide range of voltages and frequencies.



The second problem turned out to not be a problem at all. The switching phase angle appeared to change when the pulse density was adjusted, but this was not what was actually happening. As I was adjusting the pulse density, the primary current changed, and since this current is what commutates the midpoint voltage, it appeared the same way as adjusting the switching phase angle. Now that I understand it, I don't worry about it. Even if adjusted for perfect ZVS at maximum power only, the lower conduction losses will compensate for the increased switching losses at lower pulse densities. It's also possible to inject the sensed primary current into the phase setpoint to compensate, but I don't think it's worth the effort for a marginal increase in efficiency.

Feeling relatively comfortable about the design, I finalized my four layer board and ordered boards. There have been significant changes since the last screenshots I posted. The FR4 thickness between layers 1/2 and 3/4 is significantly thinner than between 2/3, so I moved around some layers to minimize bridge midpoint capacitance. I've moved the IGBTs and changed how they will be mounted. Some hold-down brackets will be glued to the bottom of the PCB, and these will clamp the IGBTs to the heatsink with even pressure over the die. Whether this is practical remains to be seen.



To modify my previous design to work like the latest version only needs some minor mods. I can try to draw up a schematic of what to do in the near future.

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Re: Adventures in induction heating
« Reply #10 on: August 28, 2017, 10:21:51 PM »
I am now through with and until the February 2015 posts and I can see a lot similarities to the problems I had with the poor mans QCW experiment, since that was exactly a quasi-resonant inverter without proper mains voltage tracking so it was unstable, glitchy and has large problems with switching transients, but that was generally also a horrible implementation when taking all your considerations into account.

It is also in the same league of problems I am facing/will face in the IKEA 2kW induction heater project, there is very little to do with in a already maxed out quasi-resonant inverter, that has a much more narrow bandwidth than your design goals. I will not say that I have given up upfront, but after more reading I am getting closer to accepting that its properly not going to give any good results...

It was great to get some experiences on this subject to take on to my own, now, more reading of the next updates :)
http://www.kaizerpowerelectronics.dk - Tesla coils, high voltage, pulse power, audio and general electronics

High Voltage Forum

Re: Adventures in induction heating
« Reply #10 on: August 28, 2017, 10:21:51 PM »

 


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