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Messages - davekni

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21
For grounding, I'll emphasize a line from my last answer: "  (Of course, for the bridge, this works only when using a bench supply or transformer, not when powered directly from line voltage.  Once running on line voltage, don't ground the negative side of C12, and don't clip the scope probe ground to anything on the bridge circuitry.  Leave the driver grounded even with direct line voltage on the bridge.)"

As long as you are running from your hy3005 supply or a transformer, ground the negative side of C12, and clip the scope probe ground to that node as well.  The ground terminal of your hy3005 supply is a good place to ground the bridge.  Don't do that once running directly from line power!  For scoping any circuit, it's best to have the scope probe ground clip connected to the ground node of the circuit you are probing.  If the circuit doesn't have a ground node, such as the h-bridge when connected to line voltage, then a differential probe is needed.  (Floating the scope ground can work instead, but only when the probe's ground lead is connected to a low-impedance circuit node, such as the negative side of C12, where injected noise won't be a problem.  Floating scope ground will now have rectified line voltage, so is a shock hazard.  And the other scope probe's ground clip will be at the same line voltage.  So, don't scope any other signals at the same time - only one signal when using floating ground.  In general, floating scope ground is risky, so I recommend not floating your scope ground.)

Yes, differential probes are useful for scoping many things, such as speaker terminals as you mentioned.  Old amplifiers usually had one speaker terminal at ground and drove the other with the audio signal.  Many amplifiers today drive both sides of the speaker with opposite-polarity audio signals.  So you shouldn't connect either speaker terminal to the scope ground.  For most audio amplifiers, the speaker signal on either side is sufficient for probing, because the other side will be an inverted version of the same waveform.  If you need to scope across the speaker, either use a differential probe, or use both scope probes and the waveform-subtract math function of the scope.  That does the same thing as a differential probe, although not quite as accurately as a good differential probe.

The H-Bridge output should be a square wave with a frequency of 240kHz (your resonant frequency).  It's always connected to either the negative side of C12 or to the positive side of C12.  The phase information I'm interested in is when the square wave edges are timed relative to the secondary high-voltage sine wave.

When the bridge is powered directly from line voltage, then there will be some rectified 50Hz sine wave signal on top of the much higher frequency square wave signal.  Since the frequencies are so different, scoping the bridge output can be useful even when line-powered.  (The scope probe ground needs to be left unconnected when probing a line-powered bridge!)

Sounds like the high-frequency signals of your SSTC are confusing the power supply circuitry.  Connecting the supply negative output to the supply ground terminal may fix that issue, making the supply behave correctly.

If I missed answering anything, please ask again.

22
Yes, the radio station thought was unlikely for you.  An AM station antenna would be large, so obvious to see.  It was just childhood memories for me.

As long as it's only the "ground" nodes being grounded, not literally "everything", then nothing will blow up.  The scope should be grounded, the driver ground (pins 4 and 5 of the UCC chips etc.) should be grounded, and the bridge negative supply (negative side of C12) should be grounded.  Then the scope probe ground can be clipped to either the driver or bridge grounds for measuring voltages.  You cannot connect the scope across two non-grounded nodes, such as measuring the voltage across C4 of the driver, because that would be grounding one side of C4.  Always measure voltages relative to ground, with the scope probe ground clip on one of the ground nodes.  (Of course, for the bridge, this works only when using a bench supply or transformer, not when powered directly from line voltage.  Once running on line voltage, don't ground the negative side of C12, and don't clip the scope probe ground to anything on the bridge circuitry.  Leave the driver grounded even with direct line voltage on the bridge.)

It sounds to me like your power supply has an internal failure causing the 44V jump.  Might still be usable though.  If its current limit still functions, that would add a bit of protection over using a transformer directly, even if the voltage isn't fully adjustable.  The supply's internal failure might make current limit non-functional as well.

Yes, higher coupling is generally good for an SSTC, until the point where the primary gets so close to the secondary that voltage starts arcs across the gap.  Look in the dark at the bottom of the secondary and at the primary to see if there's significant corona discharge.  That indicates you are close to a problem there.

As you've found, low coupling doesn't generate enough secondary voltage to couple into the antenna and start oscillation (or lock oscillation to the resonant frequency in this case with self-oscillation).

For phase measurement, here's what I was asking in reply 24:  "Coarse phasing (180 degrees or not) is easiest by trying both ways to see which one locks frequency.  Once the coarse phasing is correct and the coil is running with feedback, I suggest measuring the more subtle phase shift.  Scope the H-Bridge output with one probe and use the other probe as an antenna - just hanging in the air somewhere around the coil.  Ideally the H-Bridge output switches at or just before the peaks of the top-load voltage (which the floating scope probe is picking up).  Leave the floating probe separate from the feedback antenna to avoid changing behavior."

Now that you are running, phase measurement isn't critical.  I am personally interested in the phase measurement, however, not having any experience myself with antenna feedback.  (Antennas seem to likely to pick up other stray signals, so I've always used current feedback.)  Thank you for your willingness to provide the traces!

23
That's lots of symptoms.  I certainly won't be able to tell remotely exactly what happened.

"Floating" scopes or other equipment aren't completely floating.  There's always capacitance to the line neutral and hot wires in the scope's power supply.  The scope is likely injecting more noise when floating.  I recommend keeping your scope grounded, and keeping the driver circuit grounded.  During this initial bring-up with a bench power supply, ground the bridge negative supply as well.  With ungrounded circuitry, part of the signal the antenna sees is the noise on its local "ground" reference, the negative driver supply terminal.

(Do you happen to live close to a commercial radio broadcaster?  As a kid we lived about 1km from an AM radio station.  Any sort of antenna picked up obvious amounts of that ~1MHz signal.)

Especially without the secondary in place, the antenna may be picking up noise from any source, including floating "grounds".  The most problematic may be if the gate wiring is long enough and/or close enough to the antenna that gate-drive becomes the feedback.  There's nothing in that circuit to prevent oscillation at high frequency, which would likely happen if picking up feedback from the gate drive wires.  If you ground everything, at least that eliminates some of the noise sources, so it would be easier to see any high-frequency oscillation.

During normal operation, the high secondary voltage dominates over any other noise sources the antenna may pick up.  Testing without the secondary can still be possible as long as things are grounded and the gate-drive wiring is shielded from the antenna.

Concerning 44V, that may be a digital meter getting confused by high-frequency noise.  Cheap meters have little shielding internally.  Usually I see random quickly-changing values in such circumstances rather than voltage errors, but I have occasionally seen just an offset as you are seeing.

24
Electronic Circuits / Re: TVS diode selection for 400v transistor
« on: March 28, 2020, 04:44:17 AM »
Yes, worst-case specifications for TVS diodes don't allow tight clamping.  I've purchased quite a range of TVS diodes, finding them handy to have for my personal stock.  All have been quite close to their nominal voltage at low current and room temperature.  So, a P6KE300 is likely to have a breakdown right around 300V.  However, that voltage will depend on current, and especially on temperature.  The clamping voltage and current is specified with a total energy that heats the diode to its maximum 150C or 175C.  As with most silicon avalanche-breakdown voltages, it goes up with temperature.  So, the P6KE300's worst-case clamping voltage of 414V might not quite protect your 400V FET.  That depends heavily on the TVS current pulse duration and repeat frequency - in other words how hot the TVS diode gets.  Also, if your FET is running hot, it's breakdown voltage will be above 400V.  And, some FETs have some avalanche energy capability.  Overall, P6KE300 is probably the best choice.  Definitely not P6KE350.  Use the P6KE250 if you want to be extra safe with clamping voltage.  Still, be careful about letting the TVS do too much clamping, keeping it under its average power dissipation rating.  Or, go to 1.5KE300 for more power and higher peak current capability.

25
Are your fast diodes on heat sinks along with the IGBTs?  If not, are you sure the diodes are all still good?  I'm wondering if the diodes got hot enough to either fry or have extreme leakage current and slow switching to the point that it fried IGBTs without the diodes themselves failing.  Skip-pulse mode puts more heat into the diodes, as does secondary arcs.

Yes, DRSSTCs usually run at the lower of the dual resonance peaks of the coupled resonators, so a shorted secondary will raise the frequency.  The frequency change generally isn't significant enough to cause problems, but might be if your phase-shift circuit is particularly sensitive to frequency.  Usually the more significant effect is the increased Q of the primary, since the secondary no longer uses much power.  That causes the primary current to ramp up relatively quickly and ramp down slowly.  The pulse-skip mode will then need to skip lots of pulses, moving more power to the diodes, and likely causing more issues with gate-drive low-frequency components.  Even without pulse-skip, there will be more power for the diodes during the decay after the enable pulse ends.  (I'm working on a circuit to detect ground arcs and terminate the enable pulse before current ramps up too much.  My IGBTs have internal diodes, but their power dissipation capability is much lower than the IGBTs themselves.)

The other possible issue with external diodes is inductive voltage drop between the IGBTs and diodes.  I don't have any personal experience with external IGBT diodes.  I think IGBTs are usually rated for ~20V reverse Vce.  I don't know how sensitive they may be to spikes beyond 20V.

The "fuzzy" sine wave was likely multiple captures as the primary current ramps up.  The key reason to look there is for accurate phase adjustment.  Typically you can see the sine wave along with small steps at the top and bottom when the H-Bridge switches.  Those steps should be just barely before the top and bottom crests of the sine wave.  Try triggering on the enable pulse to get a stable waveform, or set the scope's trigger hold-off longer to prevent multiple triggers per enable pulse.

26
Yes, the fine phasing isn't that critical for SSTCs using FETs for the bridge.  It is critical for high-current DRSSTCs using IGBTs.  Still, efficiency is better with good phasing, and bridge layout (low inductance) is less critical with good phasing.

The gate-drive waveforms do look a lot like scope probe mis-adjustment.  However, the time scale is wrong for scope probe compensation.  I think your traces are likely accurate plots of UCC chip outputs.  They change rapidly until the voltage where they can't supply any more current to the gate series resistors.  Then they finish slewing as the gate capacitance is charged.

The bus supply should draw some current even if not sync'ed given the self-oscillation resistor addition - more than 18mA.  When sync'ed, it will increase several times.  At 15Vbus, the self-oscillation frequency will likely need to be adjusted closer to resonance (closer to 240kHz) to get sync'ing.  It should be possible to sync at 15V with good self-oscillation frequency.

The only reason I can imagine for FET frying at 30V is that there was some spike in gate-drive that over-voltaged the gates.  It's easy to get 24V from that gate-drive circuit, but that's generally not enough to fry FETs.  It would require a series resonance of the 0.1uF and gate-drive transformer inductance to go over 24V.  Perhaps somehow the antenna picked up the gate voltage and resonated at that frequency.  You could add a bidirectional TVS diode or back-to-back zener diodes from gate-to-emitter on each FET.  That would protect them from such gate-drive resonances.

Beyond adding FET gate voltage clamps, I don't know what other option you have other than starting to power it up.  I'd start with the current limit set low, perhaps 0.5 to 1A, just to be extra cautious.

Good luck!

27
Overcurrent is a possibility.  Some designs check only positive half-cycles, so it might be the end of the second half-cycle above 500A before shutdown.  Pulse-skipping could make that worse, as you're then hitting that current limit every few cycles, rather than once per enable pulse.

Pulse-skipping may be risky for other reasons too.  Depending on how it is implemented, it can make IGBT power sharing uneaven.  It also will make lower-frequency components to the gate-drive signal, which may require higher inductance and volt-second rating of the gate-drive transformer.  Or, if it puts some gates into the off 0V state, then there's extra half-voltage gate transitions, which are slower turn-off and may have some momentary weak turn-on overshoot pulses.

In short, I'd disable pulse-skipping mode until you have a chance to examine it closely with your scope at lower bus voltage and lower current limit settings.

The mains ground is probably fine.  I prefer to use both the mains ground and an aluminum foil (or screen or plate) ground plane under the coil against the floor or ground.

Additional 1000uF near the H-Bridge would be good.  Or, use lower inductance wiring from your existing bulk capacitors.  Multiple twisted pairs is good - that's what I needed to do even for a fairly short run.  Each pair has one wire VBus+ and one wire VBus-.  Even if you go with an additional 1000uF closer, I'd recommend twisted pairs between it and the H-Bridge.

The IGBT part you listed doesn't have an internal anti-parallel diode, so isn't a good choice for H-Bridge use.

28
That's disappointing.  Hopefully the fried IGBTs didn't take out anything else.

Was the arc from the top-load to the ground rail, or from some point part way up the secondary?  The latter is the issue I had when my secondary started 50mm below the primary.  Raised the secondary to only 25mm below and added more polyurethane coats to the secondary.  Yes, lowering the grounded strike rail can help, but keep it slightly above the primary coil.

Two possibilities come to mind.  First is that the high-frequency noise generated by a ground strike coupled into circuitry, perhaps confusing the driver, making gate voltage transitions away from the current zero-crossing points.  Besides generally keeping wires short etc., make sure the ground connection wire for the strike rail doesn't pass close to drive circuitry.

The other is what I mentioned way back in reply 25 on March 1st:  Last I saw, you have relatively long and separated wires from your bulk caps to the H-Bridge.  That inductance can cause problems at the end of an enable pulse, when the H-Bridge changes from pulling current from the bulk caps to pushing it back as it removes energy from the primary tank circuit.  That issue forced me to redo my bulk-to-bridge wiring to reduce inductance.

29
Nice to see!  All the frustrations of the process fade when things work in the end.

30
Your 60V 5A supply should be perfect for bring up this SSTC!

Coarse phasing (180 degrees or not) is easiest by trying both ways to see which one locks frequency.  Once the coarse phasing is correct and the coil is running with feedback, I suggest measuring the more subtle phase shift.  Scope the H-Bridge output with one probe and use the other probe as an antenna - just hanging in the air somewhere around the coil.  Ideally the H-Bridge output switches at or just before the peaks of the top-load voltage (which the floating scope probe is picking up).  Leave the floating probe separate from the feedback antenna to avoid changing behavior.

Not having any personal experience with antenna feedback, it seems to me that it will end up with more phase lag than ideal, with the H-Bridge output switching after the top-load voltage peak time.  That may still be fine as long as it's within 30 degrees or so.  Phasing might be better with the self-oscillation frequency set at or slightly above resonance - opposite what I'd said earlier.  Since I haven't built an antenna system myself, you'll need to experiment.  Try 150k or even 130k to see what it does to phasing.  At 60V and a 5A limit, your H-Bridge parts shouldn't be at much risk due to any phasing errors.

Concerning the UCC output waveform shape, my guess is that the upper (green trace) UCC chip happens to be slightly faster (shorter delay time) than the other UCC chip.  It switches with a fast edge, then is pulled back a bit by the increased load current when the other UCC chip switches.  As to one being warmer, I don't have any new guesses.  55C doesn't sound problematic.  Is that with the heatsink you added?  Temperature will go up a bit further once the frequency locks to 240kHz.

31
I'll take up-side-down images over the links.  It's not hard to invert them during viewing.

I agree that the bus voltage ring is not likely the issue, at least not the lower-frequency peak that I had problems with.  Thank you for the image;  I now see that the bulk cap is bolted to your bus bars, not at the other end of the cables as I was initially thinking.  That is good.

It's hard to tell phase lead based on the attached scope traces.  H-Bridge output phase appears to be roughly at the current zero-crossings.  Slightly before current zero-crossing is ideal, so slightly more phase lead.  More phase lead requires a larger inductance on the driver board.  (I have UD2.7 schematics, so am presuming that UD2.5 is similar.)  Hopefully phasing the H-Bridge output slightly before current zero-crossing will reduce the spikes.  It usually does.  It is possible that those high-frequency spikes are even higher voltage at the IGBT die themselves than where you can scope.  If the spikes can be reduced by adding a bit more phase lead, that might make the IGBTs last longer.

32
Depending on the frequency and thickness of the graphite crucible, the magnetic field may be mostly blocked from the aluminum, all dissipated within the graphite.  In that case, properties of the inside material don't matter.

Even if the crucible is providing all the heating, I believe it will take much more than 1kW to melt 500g of aluminum.

33
Great!  Adding the resistor actually increases sensitivity to antenna feedback, as it charges the antenna voltage most of the way for each half-cycle, so only a little extra charge from the antenna is needed for each transition.  Once the secondary voltage ramps up, the antenna feedback current will be much larger than the resistor current, so the resistor will have little effect.

It's not surprising that the UCC chips warm up, as they are running continuously now.  No reason I know for one to heat more than the other.  Likely one happens to have a bit higher drive strength so runs a bit cooler.  The only load difference would be parasitic wiring capacitance, not significant for heating.  Heat-sinking is a good idea, since most SSTC designs don't run at 100% duty cycle.  You won't be able to use an interrupter without the enable pins working.

There are many variations for grounding.  My preference is to lay aluminum foil or sheets or screen on the floor/ground, then wire the secondary ground to the foil and to the power line ground.  Grounding details are more important for SGTCs and DRSSTCs, as they have high peak currents.  You should be fine with just the line ground wire for an SSTC.  (Thinking back, that's all I used for my SSTC.  Haven't had that out in years.)

I've always used current feedback, so hopefully you've found other information on just how close to place the antenna to the secondary to be safe from arcing over.  To start up, place the antenna near the secondary however is typical, apply gate-drive power, then initially some low half-bridge power, perhaps 30V.  With the self-oscillation, 30V bus is likely enough to get it to sync. up.  You can see with the scope if gate-drive (or half-bridge secondary) frequency jumps up to 240kHz.  If not, you can try raising the bus voltage further, or drop the self-oscillation resistor to ~160k to get closer to 240kHz starting point.

Do you have any current-limiting in your bus supply?  A current-limited bench supply would be great for start-up testing.  If running directly from a variac, adding an incandescent light bulb in series between the variac and rectifier bridge is a good current-limit option.  The half-bridge current will be low until the antenna feedback locks the frequency.  Current will jump up when it locks.  Running continuously may draw too much current to allow reaching your intended bus voltage before current gets too high.  Check gate-drive phasing before approaching full power.

34
I'll agree that counterfeit Chinese parts are the most likely cause.  As you mentioned, drive phasing is another possible cause.  That you can measure with your scope.

One small point that isn't likely to be a real problem:  Those IGBT packages are intended to have one emitter terminal used for the power connection and the other for gate return.  It's better not to have connections between the two emitter terminals outside of the IGBT package itself.  Outside connections add some of the voltage drop (due to parasitic inductance) to the gate-drive signal.  Your connections are with aluminum bars directly on the packages, so not likely an issue.  I'd still suggest leaving off those little bars.

I couldn't tell from the images just how much wiring inductance there may be between the 2.5uF snubber cap and the bulk filter cap.  With 600V IGBTs, inductance there can be problematic at the end of each enable pulse, when the H-Bridge changes from drawing power to sourcing power (from the Tesla primary back to the bulk supply).  I had to redo my bulk-cap wiring for that reason, as my DRSSTC also uses 600V parts (10x TO247 for each leg).  Bulk cap inductance makes a large voltage spike on the snubber at that end-of-enable transition.  Try scoping across the snubber and triggering off the trailing enable edges.

Final note:  For future posts, would you mind uploading pictures directly to this forum rather than linking?  The linked image host has intrusive advertising, and was initially blocked by my anti-virus software.

35
The 0.1uF capacitor is in the gate drive circuit.  The relevant capacitance here is U2-1, the antenna connection.  CD40106 data sheet lists 5pF typical input capacitance.  I can't find capacitance curves for 1N60 diodes.  The antenna may add another 5pF - depends on how long it is and what other conductive parts it gets close to.  So, the capacitance to use in calculating self-oscillation is around 10 to 15pF.  It is just the first schmitt-trigger inverter stage involved in the self-oscillation, so only U2-1 and U2-2.  You can test self-oscillation without anything else powered up.

36
Electronic Circuits / Re: Smoking power resistor wattage overhead
« on: March 26, 2020, 12:09:02 AM »
I doubt it would get into the uH range at 22 ohms.  Doesn't take that much wire unless resistance is well higher.  If I had access to the nice impedance bridge at work, I could quickly measure a few wire-wound resistors near that resistance.  But, social-distancing measures prevent that for now.

It's not quite the same as leakage inductance, as it's included in the sense voltage.  Will just produce a bit of phase-lead.

37
Electronic Circuits / Re: Smoking power resistor wattage overhead
« on: March 25, 2020, 10:01:13 PM »
When you say "As expected it runs red hot", is that literal?  If it's actually glowing, I suspect it's dissipating more than 2.5 watts.

Power resistors, especially wire-wound ones, are generally designed to run quite hot at rated power.  If you get organic contamination on the surface, smoke isn't surprising.  But that should burn off and the smoking stop after a few minutes.  I like to have some margin below spec limits, but regularly use resistors above 50% of rated power.

Does your design mind the inductance of a wire-wound resistor as in the 7 watt link you shared?

Your image of 1.8 ohm 2 watt resistors looks extremely close to bags of resistors I purchased surplus years ago - same values and appearance, with only slight differences in marking.  I think these are film resistors, not wire-wound.  The inductance is low for wire-wound, around 15nH if I recall my measurements correctly.

38
Agreed, you must have bad UUC chips :-\  That's an unusual failure, not the typical fried output stage from overload.

For the resistor addition to make it self-oscillating, try searching for "schmitt trigger oscillator".  Here's one good page:
http://electronics-course.com/schmitt-trigger-oscillator

For this driver, the input-to-ground capacitor is just the parasitic capacitance of the input (U2-1) and the parts connected to it, the two clamp diodes (D1 and D2), and the antenna's capacitance.  The added resistor from U2-1 to U2-2 controls how fast the capacitance is charged and discharged.  If the added resistor is large, the frequency will be low, as the input capacitance will be charged and discharged slowly by the low current.  A smaller resistance will charge and discharge the capacitance faster, resulting in a higher frequency.

If this self-oscillation frequency is close to resonance, then the Tesla coil secondary will build amplitude for several cycles before getting too far out-of-phase.  That provides more amplitude for the antenna to pick up, so makes startup more certain.

39
I'm wondering if there's a cold solder joint in the enable signal path.  Have you tried probing on the U3-3 and U4-3 pins themselves, up where the pin enters the epoxy package molding?  Or, use an ohm-meter to check continuity between U3-3 and U2-4, with both probes up on the IC package adjacent the black epoxy.  The UCC parts have an internal pull-up resistor on enable, so a disconnected pin will behave as enabled as you are seeing.  My only other thought for such behavior is that the U3-3 and U4-3 experienced some mechanical stress that broke the wire bonds inside the chips.  That seems unlikely - usually requires aggressive bending of pins during desoldering and resoldering of parts.

Adding the 1-2meg resistor across U2-1 to U2-2 should allow continuous operation with enable behaving as permanently true.  Low bus voltage is a good idea for such continuous running.  If you know your secondary resonant frequency at least roughly, it's ideal to have the self-oscillation frequency from the added resistor match the resonant frequency.  You could adjust the resistor value to get that result, lower resistance for higher frequency.  It's better not to get the self-oscillation frequency higher than resonance, so err on the low side.  The 1-2meg will likely be on the low side - why I suggested that range.  Have the antenna attached for this adjustment, but no bus voltage.

Until the UCC enable issue is fixed, you will be limited to continuous operation.  It would be possible to patch with gates, but that's quite a nuisance.  One driver chip is inverting, so needs to have its input pulled high for disable.  The other is non-inverting, so needs to have its input pulled low for disable.  Hopefully it's just a bad solder joint and can be easily fixed.

It's difficult to guess what happened when you touched the antenna, and what signals your body was injecting into U2-1.  In many situations, the dominant signal from a person touching a scope probe is line-frequency, 50 or 60Hz.  Low-frequency switching will generate 24V gate voltage pulses because C4 will charge to 12V then to -12V.  The FETs are probably specified for 20V maximum Vgs, but not likely to fry at 24V.  If the signal injected from your body happened to hit the series-resonance of C4 with the gate-drive transformer, then even higher gate voltages could be generated.

40
FF400R12KE3 is specified at 800A pulse current (at 1ms), so that should be plenty safe.  Looking at the typical plots, there appears to be quite a bit of headroom past 800A, especially if the gate voltage is above the specified 15V.  Hopefully others here will offer advice from their more extensive experience.  My limited info from other IGBT bricks is that they tend to fry at about 2x their pulse rating or 4x their continuous current rating, which would be 1600A for this part.  I'd suggest staying well below 1600A.

Inductance of your bus voltage routing, from the IGBT parts to snubber and bulk capacitance can be limiting too, if transients exceed 1200V.  A good layout with low inductance should keep internal IGBT voltages well below 1200V even up to 1600A.

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[Electronic Circuits]
davekni
April 02, 2020, 08:20:25 PM
post Re: (DR)SSTC music MIDI files
[Dual Resonant Solid State Tesla coils (DRSSTC)]
DJthefirst
April 02, 2020, 08:20:07 PM
post Re: Power MOSFETs with fast recovery body diodes
[Electronic Circuits]
davekni
April 02, 2020, 08:09:29 PM
post Re: A few (stupidly basic) questions
[Beginners]
Mads Barnkob
April 02, 2020, 02:53:24 PM
post Re: A few (stupidly basic) questions
[Beginners]
John123
April 02, 2020, 11:46:07 AM
post Re: TVS diode selection for 400v transistor
[Electronic Circuits]
John123
April 02, 2020, 11:40:55 AM
post Re: Power MOSFETs with fast recovery body diodes
[Electronic Circuits]
John123
April 02, 2020, 11:32:19 AM
post Re: GDT keeps on killing IGBTs
[Dual Resonant Solid State Tesla coils (DRSSTC)]
ritaismyconscience
April 02, 2020, 03:48:55 AM
post Re: Class E SSTC Topology
[Solid State Tesla Coils (SSTC)]
ZakW
April 01, 2020, 09:47:42 PM
post Re: GDT keeps on killing IGBTs
[Dual Resonant Solid State Tesla coils (DRSSTC)]
davekni
April 01, 2020, 09:31:49 PM
post Re: Power MOSFETs with fast recovery body diodes
[Electronic Circuits]
davekni
April 01, 2020, 08:47:11 PM
post Re: Fishing for atmospheric electricity
[Static Electricity]
johnf
April 01, 2020, 08:45:14 PM
post Re: Homemade HV transformer
[Transformer (Ferrite Core)]
johnf
April 01, 2020, 08:37:29 PM
post Re: TVS diode selection for 400v transistor
[Electronic Circuits]
davekni
April 01, 2020, 08:24:06 PM
post Re: Pt100 oddities
[Electronic Circuits]
davekni
April 01, 2020, 08:16:59 PM
post Re: Pt100 oddities
[Electronic Circuits]
johnf
April 01, 2020, 07:50:16 PM
post Re: Fishing for atmospheric electricity
[Static Electricity]
haversin
April 01, 2020, 07:42:47 PM
post Re: Homemade HV transformer
[Transformer (Ferrite Core)]
John123
April 01, 2020, 07:39:48 PM
post Re: Fishing for atmospheric electricity
[Static Electricity]
John123
April 01, 2020, 06:07:44 PM
post Photographing/filming high voltage arcs
[DSLR]
John123
April 01, 2020, 05:17:04 PM
post Power MOSFETs with fast recovery body diodes
[Electronic Circuits]
John123
April 01, 2020, 02:01:30 PM
post Re: Pt100 oddities
[Electronic Circuits]
kamelryttarn
April 01, 2020, 01:38:47 PM
post Re: Pt100 oddities
[Electronic Circuits]
T3sl4co1l
April 01, 2020, 12:49:21 PM
post Re: TVS diode selection for 400v transistor
[Electronic Circuits]
John123
April 01, 2020, 10:29:06 AM

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