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Messages - davekni

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1
Yes, for 500A and ~150kHz, 100nF MMC is probably better.  Hopefully that will increase your arc length.

2
There is something very odd about the current traces after shut-down.  I'm doubtful that the driver is shutting down due to normal over-current, especially since the LED doesn't light.  Perhaps it's something about the IGBTs getting too hot to function, but without permanent failure.  Probably incremental damage that will eventually lead to failure.  Silicon die can often survive 275C for short periods IF the resulting thermal-expansion mechanical stress doesn't crack die-attach or bond wire connections.  Perhaps your IGBTs are packaged in a way that survives such internal temperatures.  The biggest surprise to me is that the IGBTs at that temperature aren't increasing conductivity in the off-state to the point where they go into thermal runaway.  I don't have experience with hot IGBT die, however, rather with ~60V ASICs that I've designed and tested to failure.  Perhaps the hot IGBTs are somehow shorting-out the gate-drive signal, or not turning on even when it is high.

3
Yes, I was thinking about that last night.  It's not directly stealing current.  Rather, the phase-lead circuit is increasing the load impedance on that current sense output, which is likely enough to cause the large first-stage current-sense transformer core to saturate, which is reducing signal to the over-current second-stage.  Given your phase-lead patches, it's probably best to make completely separate current-sense transformers for feedback and for over-current.

4
Beginners / Re: A few (stupidly basic) questions
« on: April 02, 2020, 08:58:12 PM »
Mads,

Isn't SSTC frequency determined by the secondary and top-load?  So the 250kHz example in your design guide would remain constant, not go up to 1MHz.

One point I forgot to mention for scaling voltage: primary geometry.  To scale properly, the primary winding needs to remain the same size.  For scaling from 320V to 80V, the primary in your example needs to be four paralleled 2-turn windings to fill up the same vertical height.  To keep wiring inductance low enough, each two-turn winding would need to route down to the H-bridge separately, ideally as four twisted pairs.

The above primary winding should have 1/16th of the inductance (same geometry, 1/4 turn count), or 635nH, for 1.0 ohms at 250kHz.  That's 80 amps at 80 volts, for the same reactive power as 20 amps at 320V.

For the above comparison, I've used the calculations as they are on your SSTC design guide.  However, the actual peak currents are higher, since square waves have more area than sine waves.  The current through an inductor with square-wave voltage is a triangle wave.  For 250kHz and 320V, the coil sees +320V for 2us.  That ramps the current through a 10uH inductor from -32A to +32A (640uVs).  The peak current is 32A.  Of course, this is with ideal assumptions - no coil resistance and no H-Bridge switching transition time.

Separately, I like the nice picture in your SSTC design guide showing the tight coupling of primary coil to the bottom of the secondary coil.  That prevents that lower secondary portion from building up even higher voltage due to resonant current, since that voltage is tightly coupled to the primary's fixed voltage.  (I had been wondering about how the physical proximity didn't cause breakdown issues, until realizing that it also clamped the lower secondary voltage.)

5
Electronic Circuits / Re: TVS diode selection for 400v transistor
« on: April 02, 2020, 08:20:25 PM »
John,

Probably best to switch to a part that is working well.  Turn-off speed is certainly important for flyback use.  The MJE13007 inductive turn-off characteristics are specified with -2.5A base drive.  With lower base-drive currents, other transistors designed for higher-frequency might function better.  (My guess as to why -5V is better for this part is that internal base resistance requires that much voltage to get the -2.5A recommended for inductive load turn-off.)

I don't know if the beta degradation has a plateau or not.  In general it's a cumulative process, so more use should make it worse.  Since the goal is usually to avoid degradation, data on extended degradation probably isn't valuable enough to measure.

6
Electronic Circuits / Re: Power MOSFETs with fast recovery body diodes
« on: April 02, 2020, 08:09:29 PM »
John,

Yes, I never rely on avalanche ratings either.  However, my limited and unscientific observations would suggest that parts with higher avalanche ratings tend to be more robust.  Perhaps that's due to occasional unintended over-voltage spikes during startup current peaks combined with parasitic wiring inductances.  The reduction with SIHG039N60EF is small, so I'd take the fast diode over the small avalanche energy difference.

That does look like a very nice FET you've found.  That's certainly one I'll consider if designing another high-power FET circuit.  The only caution I have is that the low Crss makes for very rapid slew rate on Vds.  I have some 200V FETs with similar on-resistance and low Crss.  They kept frying in an H-Bridge design with transformer-coupled gate drive because the high slew rate produced gate currents due to winding-to-winding capacitance that were so high frequency that the leakage inductance of the transformer wasn't low enough to shunt them.  That feedback caused a couple cycles of very-high frequency full-amplitude oscillation at each intended H-Bridge switching point.

7
Phase lead looks great now!  Double-check that when you get back to full current, as phase lead may be current-dependent given the driver's clamped feedback load.

If I recall correctly, you are using 10 ohms on your scope current transformer.  The driver's over-current looks more like 1 ohm plus two diode drops.  Perhaps that's the scaling issue - a different driver load resistance than what you are using in calculations.

8
Electronic Circuits / Re: Power MOSFETs with fast recovery body diodes
« on: April 01, 2020, 08:47:11 PM »
Usually there's a trade-off that faster diodes have higher forward voltage drop.  To my surprise, that isn't the case for these parts.  I did notice that the avalanche energy rating was a little lower for the fast-diode FET - no idea why.

9
Electronic Circuits / Re: TVS diode selection for 400v transistor
« on: April 01, 2020, 08:24:06 PM »
Do you have any MJE13007 parts you haven't used yet?  You could try it at -5V instead of -9Vbe.  The beta degradation is tied to reverse BE leakage current, which typically rises sharply as you approach the -9Vbe spec limit.  I don't know if switching speeds or any other parameters degrade with beta.  You might find that an MJE13007 part operated within recommended conditions works well.

Yes, MJE13007 is an unusual part relative to my experience.

10
Electronic Circuits / Re: Pt100 oddities
« on: April 01, 2020, 08:16:59 PM »
If I'm understanding correctly, you are down to calibrating the cheap Pt100 sensors, as your electronics are repeatable.  If so, a scale factor is more appropriate than an offset.  Being cheap, these are likely thin-film RTDs, perhaps rejects where the deposited platinum was too thin so couldn't be laser-trimmed to 100 ohms at 0C.  Measure the actual resistance at 0C.  Calculate the required sale factor to get to 100 ohms.  Then multiply all resistance readings by that scale factor prior to converting to temperature.

I think platinum resistance/temperature curve extrapolates to 0 ohms at -260C.  So, if your 4-20mA results are already in degree C units, add 260, multiply by the scale factor, then subtract 260.

11
Are the sine-wave traces of the primary voltage, the junction between the MMC and primary coil?  Presuming so, it looks like a bit too much phase lead, although that's definitely better than phase lag.  Ideally the H-Bridge voltage steps on top of the primary voltage would be just 5-10 degrees or so before the peak.  Looks like perhaps 25 degrees now.  Would need to zoom in closer to tell.

Yes, the low voltage makes IGBT capacitance higher, so ringing tends to be worse.  Lower current may change the behavior of your phase-lead circuit, since the driver board's input impedance isn't linear with voltage.

12
Beginners / Re: A few (stupidly basic) questions
« on: April 01, 2020, 01:32:54 AM »
Mads,

I'm curious about your line: "I have only built mains powered SSTCs as I find the low voltage versions to be harder to get to work and they also seem to have lower spark-length-efficiency, I am a sucker for long sparks".  If the power levels are equivalent and the part impedances are all scaled appropriately (larger capacitors, lower inductances, etc.), what makes the difference?  The best guess I'd have is that wiring inductance through the primary circuit (H-Bridge to primary coil) isn't reduced enough.  Low impedance would require twisted pairs and tight layout.  Do you have other possible reasons?

Thank you.

13
Thank you for the scope traces!  Do you know if both gate-drive outputs of your driver board always switch together?  I'm guessing so from the traces, but it's a bit hard to tell.  Pausing only one half of the H-Bridge makes a more efficient pulse-skip mode.  It makes the H-Bridge output go to ~0, rather than to inverted, so makes the current ramp down more slowly.  Your scope traces show current ramping down fairly rapidly during pulse-skip periods, leading me to guess that the two gate-drive outputs are the same.

There are two possible issues with this form of pulse-skipping.  First is the same bulk-cap to H-Bridge wiring inductance we've discussed already.  It's now being driven multiple times in a row.  If the skip pattern happens to hit the resonance of that inductance and your local H-Bridge snubber capacitance, the local VBus voltage peaks at the bridge will get higher.  However, the one scope trace of an H-Bridge output doesn't show evidence of such a problem.  What current was running for the H-Bridge output scope trace?  It may be worth scoping an H-Bridge output more as you set current limit higher again.

The other possible issue with pulse-skipping is the positive ring on the gate-drive signal when transitioning from negative to 0V.  (For example, see the upper trace of the scope plot labeled "Left side gate drive and right side gate drive".)  If that spike is enough to momentarily turn on an IGBT, before the opposing one is off, it would cause current shoot-through.  (The IGBT turning off will do so more slowly than normal, as it's gate voltage is going from positive to 0, not to -18V.)

Yes, there does appear to be an issue with current limit scaling.  That could be the entire issue.

Phasing appears to be about at 0 degrees.  It is best to have a little bit of phase lead, so IGBTs turn off slightly before current reaches 0, allowing the remaining bit of current to cause the voltage swing before the opposing IGBTs turn on.  So, yes, slightly more phase lead would be ideal.

14
Electronic Circuits / Re: TVS diode selection for 400v transistor
« on: March 31, 2020, 04:44:05 AM »
The MJE13007 data sheet has a graph called "Maximum Reverse Bias Switching Safe Operating Area", figure 7.  It shows the allowed current at turn-off vs. Vce, with different graphs for different levels of Vbe, 0V, -2V, and -5V.  I see that it's better with more negative Vbe.  Hadn't ran into that situation before.  It may improve farther to -9Vbe, but it's also possible that the curve would reverse direction and allow lower Vce at -9Vbe.  If the Vce punch-through voltage is only slightly above the 700V avalanche-breakdown voltage for Vces, then the more negative base could bring the punch-through breakdown voltage below 700V.  So, -5Vbe is good, and -9Vbe MIGHT be better.  (Punch-through is when the reverse-biased depletion regions for C-B and B-E meet in the middle of the base region.  It's a different voltage breakdown mechanism than the more common avalanche breakdown.)

Notice that -9Vbe is the maximum allowed.  An 8.2V nominal zener plus diode forward drop will be typically about 9V.  Worst case it could be a bit higher.  For a one-up hobby project, that's likely fine.  Might drop to a 7.5V zener if making more than one.  One other consideration from my brother Dan:  Many bipolar transistors have their properties, particularly beta, degrade when operated with reverse Vbe for extended periods.  This is true especially when Vbe is close to breakdown voltage.  Dan's experience is with high-frequency BJTs, not power devices, so this may or may not be significant for this device.

That zener looks plenty capable for several hundred mA.  No need to use a TVS instead.

15
A documented driver is a good idea.  If you want to continue learning about this driver, while you have over-current set low, scope the gate-drive outputs of your driver.  It may be tricky to get all the information with a two-channel scope.  Can you get into a stable mode, perhaps w/o secondary, where the pulse-skipping is consistent from one firing to the next?  It would be interesting to see how this driver runs pulse-skip.

16
Electronic Circuits / Re: TVS diode selection for 400v transistor
« on: March 30, 2020, 06:35:13 PM »
Yes, Vces (700V) just needs the base pulled down to the emitter voltage to sink the collector-to-base leakage current.  I'm rusty on my semiconductor physics, but having the base below emitter voltage by almost Vbe breakdown might actually make things worse than 0Vbe.  I need to consult my brother, as he is more versed in such matters.  I'd suggest clamping the base voltage to one diode-drop below emitter just to be safe.  There's no value in going farther negative.

The Vces is typically a static (DC) test.  There may also be issues during turn-off, when there's still some minority carriers in the base region (base storage charge).  Again, I'll need to think about this more and/or consult my brother.

17
Beginners / Re: A few (stupidly basic) questions
« on: March 30, 2020, 02:23:23 AM »
Curtis,

Line voltage can be dangerous, but microwave oven transformers (MOTs) are MUCH more dangerous.  A man died here in Wilsonville a few years ago using an MOT for burning lichtenberg figures, accidentally touching the wires.  (I made an SRSGTC using two series-connected MOTs, 2.5kW average power.  That's my toy that scares me the most to run.)

Yes, SSTCs are generally safer.  And, yes, you can run one easily from 60V 30A - that's plenty of power.  Just requires higher current FETs or IGBTs, larger DC blocking capacitor, and fewer turns on the primary.

Can't comment much on RF regulations, which generally start at 30MHz for radiated, lower for conducted down line cord.  I suspect most Tesla Coils violate such regulations, at least when making strikes to a ground target.

If running with an interrupter, so pulsed operation, it's quite possible to have higher peak currents while keeping the average under 15A.

If you are using a bunch of scavenged parts (which is great), the first step is to measure them.

18
Mads, thank you for the compliment, and especially thank you for moderating this forum!  I was thrilled to come across it last September.

"So it looks like the exact 5 IGBTs survive each time. It's always the 4 IGBTs on the left and the one on the top right. All the other 3 fail. I think this might be because I'm using 550V TVS, and maybe the IGBTs on the left got "lucky" and got the ones that triggered on lower voltage. I should probably switch to 440V. However, nothing bad seems to have happened yesterday, and I tested it at higher voltage. Maybe the extra current contributes to higher voltage spikes?"

Yes, most spike voltages are roughly proportional to current.  IGBT capacitance is voltage-dependent, and other IGBT parameters depend on voltage and current, so it's not completely simple.  In particular, the low-frequency (wide) voltage spike at the end of the enable pulse caused by bulk-capacitor to bridge wiring inductance is quite linearly proportional to current.  That would be my first guess for this latest frying.  Uneven current sharing is also a possible cause as Mads pointed out.

Yes, reducing the TVS voltage would be a good idea.  If the voltage spike has too much energy for a lower-voltage TVS to survive, at least a fried TVS diode would provide information that there was an over-voltage event.  (Although it's also possible that an over-voltage event is secondary, caused by the first IGBT frying, before the opposing one(s) fries.)

19
Electronic Circuits / Re: TVS diode selection for 400v transistor
« on: March 29, 2020, 09:08:39 PM »
The 258V clamp voltage will increase if the diode is required to clamp repetitively, enough to get it hot.  But that is a great ratio from 250V rating to 258V clamp even for single-pulse, a good low-impedance part.

Yes, I'd go for 300V.  It isn't quite guaranteed worst-case (414V), but likely to be fine.

The best I can tell, the 1500W diodes are equivalent to 2.5 paralleled 600W diodes.

The only real down-side to bidirectional TVS diodes that I'm aware of is slower clamping in the ns range due to forward recovery time of the forward-biased diode.  Capacitance is generally lower for bidirectional TVS diodes due to being two in series.  The clamping speed should be no issue for your use, since the flyback capacitor limits voltage slew rate already.

20
Did you turn off pulse-skipping mode?  Or is that not possible?  At half inductance and the same frequency, you'll be running about twice the current, so much more likely to high current limit and pulse-skipping mode, which may fry more IGBTs.

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