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Messages - davekni

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1
Spark gap Tesla coils / Re: SGTC MK1 - An Accomplishment in Progress
« on: November 22, 2019, 06:21:22 AM »
Those temperatures seem quite reasonable.  The only issue would be if the secondary power dissipation was concentrated mostly in the diodes, which could make their internal die temperature much hotter than secondary surface temperature.  That's why I was asking about secondary current.  A wild guess for that flyback is ~5mA average (DC) secondary rated current.  You could probably push it a bit higher, especially since the sine-wave drive has the diodes conducting with a higher duty cycle than a normal flyback waveform.

Here's the ZVS I'm using to simulate something like your setup.  The voltages and currents are different, as this is derived from my Jacob's ladder ZVS.  The input ramps from 0 to 160V (most of what it sees from rectified 120V line power).  The secondary is 1:1 to the primary.  That's just for convenience in comparing input and output waveforms on the same simulation plot.  It's easy to scale the output manually.  It also uses a separate high-coupling-factor center-tapped coil to feed DC to the oscillator.  That's left over from some of my old ZVS induction circuits - works roughly the same as center-tapping the working coil (L3 in this case).  You can modify it to be like your circuit.


I wound a flyback from my stash (11 x 13.5mm rectangular core area) with 10T and made some measurements.  Coupling factor was 81%, and the secondary has somewhere around 1600 turns.  Your flyback has 74mm^2 core area, so can handle a bit more volt*seconds than my example calculation using 50mm^2.  I think you've built a very nice close-to-optimum system for these flybacks.

A resistor in series with the negative ground-return side of the secondary would allow measuring current.  1000 ohms would have 5V drop at 5mA.  The waveform will be higher pulses with zero between, which you could measure with a scope.  If using a meter to measure the voltage-drop, I'd suggest adding a 0.1~1uF capacitor across the resistor so the high-frequency pulses don't confuse the meter.  If using a scope, you can also see the spark frequency, as the current will drop as the MMC charges, then suddenly rise again after each spark.

In simulation I played with feeding the ZVS from only one side (removed L1 and L2, made L4 200uH and wired it to either VP or VN).  Moving the DC feed all the way to one side was about right to compensate for the DC component of output current.  It compensated well at the medium-to-low current conditions (higher output voltage).  At low output voltage and high output current, even that wasn't fully compensating for output current.  I'd initially guessed the "center"-tap would need to be moved only 1 or 2 turns from center.  Turns out that all the way to one end is good.  Feeding to VP should be compensating, adding current to the end of L3 that matches the positive end of L5 where current is leaving.  I'm 99% sure that's what worked in simulation, but was too distracted today to write it down.  Feeding to the wrong side doubles the total DC current bias to the core.  To check for yourself, wire it each way and plot the sum of L3 and L5 current to see the net DC component.  (Another reason it's easier to simulate with 1:1 turns ratio.)

Yes, if/when you get to DRSSTC, I'd build your own.  Much more fun and more educational too.  You're off to a great start!

2
I found some data on the first two 200V parts, which are plenty fast enough.  The voltage rating may be a problem, however, unless your IGBT bridge has very-low parasitic inductance.  I'd guess 120ns would be sufficient.  If the third option w/400V is on that order of speed or better, it should be good.  Even a bit slower may be OK.

3
Have fun!  I think Dave Kni mentioned a disk launcher in his intro a couple months ago.
My original disk-launching project was years ago, 2003.  Started with coin-shrinking in 2002, then a bit of disk launching (using the same ancient 14uF 20kV oil/paper pulse capacitor).  Disk launching was originally going to be my first HV project.  Found a few of the disk-launching artifacts in the garage:



Above is an image of an un-launched 5cm disk punched from sheet aluminum, remaining scraps of what was the 16AWG launching coil, the small plywood board that the coil was taped to (with wire impressions from the launch), and the larger target plywood with trip-wire to measure speed.  The disk folded during the launch and buried itself into the target wood.  Speed was barely over sound, 345m/s.  I think the capacitor was close to fully charged, so about 2800 Joules.



This image is of a related test - attempting to form aluminum into a shallow mold.  Didn't work well - perhaps need holes for air to escape.  The coil under the white tape was used once or twice, cracking the tape.  The other smaller coil wasn't used.

Finally, recently (2016), I made a couple small disk launchers (penny launchers).  Here's my video of the first launcher.  The completed battery-powered unit is shown in the final 30 seconds.  The first 6 minutes show simpler examples, starting with a small air-core coil under a penny with a mechanical switch, adding an iron core, changing to a TRIAC switch, etc.  That launcher used 30uF DC link capacitor charged to 550V (slightly above rated 500V), with a core ground from standard line-frequency laminated steel I-core.  I think the coil is 20 or 30 turns of 24AWG magnet wire with epoxy coating, but don't recall for sure.  Could be 27AWG.  Perhaps I'll find notes at some future point.  (The second unit, not shown, uses two 30uF caps in parallel.  It can launch old copper pennies 2~3 meters up, and aluminum disks 4~5 meters.  Haven't measured exactly.)

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4
Great to hear that it's working better!

Did you try the 1meg with an HC14, or just with the 4069?  It should help with either one, but the HC14 with resistor should self-oscillate even with no supply to the IGBTs.  If you try 1meg with HC14, measure the frequency with no IGBT power (but with the antenna connected).  Change the resistor by the ratio of measured_frequency / desired_frequency, where desired_frequency is 220kHz or whatever your updated resonant frequency may be.

Have you checked the polarity of the H-Bridge?  Does it work any better with reversed polarity (reverse gate-drive leads from driver-chips, or reverse H-Bridge output to primary coil).

Yes, I used a current-transformer in the secondary ground lead, but I wouldn't copy my specific implementation.  It is self-oscillating, and locked to the secondary in a relatively-narrow frequency range, so needed fairly precise tuning to work well.  I think the conventional circuit you have could work with a current transformer in the secondary ground lead, with a load resistor and then capacitor to HC14 pin 1.

5
Spark gap Tesla coils / Re: SGTC MK1 - An Accomplishment in Progress
« on: November 21, 2019, 05:44:40 AM »
I hadn't intended to bring up DC bias, but that is a good point.  The half-wave rectification of the internal flyback diodes will create some DC magnetic field in the core.  Core saturation will show up as distortion to the sine-wave shape of the ZVS output (flyback primary) leads.  Saturation will show up as steeper edges and flatter top compared to a true sine wave.  If one ZVS output has steeper edges than the other, that's the result of the DC field component caused by the output diodes.  Compensation could be done by moving the "center" tap a bit off of center.  Then the DC from the ZVS supply input inductor will flow through more turns one direction than the other, so add/subtract DC bias.  DC bias compensation may not be necessary depending on how close you happen to be to saturation.  I have a ZVS simulation with full-wave rectification on the output.  I'll change it to half-wave to get some feeling for the bias issue.  (Saturation isn't simulated, so it won't be perfect.)

Frequency is determined by the flyback primary inductance and your ZVS resonant capacitor value (and the secondary winding capacitance adds a bit too).  Changing frequency involves just changing ZVS capacitor value.  What value do you have now for your 70kHz result?  Did you get a chance to measure core cross-section (ferrite thickness and width inside your primary winding) - at least a guess?

Do you have any information on the average DC output current?  For example, if you know the spark repeat frequency and the spark-gap voltage (9kV last you mentioned), then the average current can be found by 16.5nF * 9kV * spark_frequency, where 16.5nF is your MMC capacitance.

You may not be far off from optimum already.  If the flyback is staying below 100C, it may be fine as is.  (I'd still suggest adding the output inductor string to protect from nano-second fall times.)  In other words, analysis and optimization is fun, but you may prefer to go with what is working and not drag out the details.  This is a fun project, but don't let me take it too far down the analytical path if that's not your interest.

6
Spark gap Tesla coils / Re: SGTC MK1 - An Accomplishment in Progress
« on: November 20, 2019, 05:30:25 AM »
Thank you for the images.  I'd be quite surprised if they don't have internal diodes.  I can't see how your SGTC would work at all if they didn't include diodes.

Should have been obvious to me, but I hadn't realized that you were winding your own primary.  Flyback transformers don't typically have center-tapped primary windings, and you mentioned center-tapped primary in your initial post.  Custom primary winding has a likely-key advantage besides center-tap:  The coupling factor will be lower than with the built-in primary winding.  As I discuss some in my ZVS Jacob's ladder post:
    https://highvoltageforum.net/index.php?topic=831.0
Having a coupling factor below 0.86 allows the ZVS oscillator to run over the full range of output loading from short to open.  That's close to what you have as the MMC charges from 0V to the spark-gap trigger voltage:  shorted load to some lower load current (higher voltage).  It would be interesting to measure the coupling coefficient of your flyback.  However, that's not necessary unless you have trouble with the oscillations dropping out, which would cause run-away input current to the ZVS oscillator.

What frequency is the ZVS running with your new small flyback transformer?  What is the cross-sectional area of the flyback's ferrite core?  Knowing those two values would allow estimating the ZVS input voltage permitted before saturating the core.  I'll use 50kHz and 50mm^2 area for an example, and a presumed saturation flux density of 0.4T.  Period = 20us, divided by 2PI = 3.183us/radian.  50mm^2 * 0.4T = 20uVs/turn, for 200uVs for  your 10-turn primary.  200uVs / 3.183us = 62.83V peak.  ZVS peak voltage is ideally (no losses) PI * DC_input_voltage.  So, input voltage would be 62.83V / PI = 20Vdc.  If the core could handle 0.5T (the highest I've seen for any fferrite material), then Vdc could be up to 25V.  Of course, this is just for my example guess for frequency and core cross-sectional area.

If you want to optimize power from the small flyback, it can help to separately measure temperatures of the primary winding, ferrite core, and secondary winding.  That indicates where to improve.  Hot primary might be improved with litz-wire.  With your single-layer primary, litz is of less advantage than for multi-layer windings.  So, I'd guess that won't be the hottest.  If the core is hottest, it could be due to flux saturation and/or high frequency losses.  A hot secondary is likely due to the internal diodes, either just total current or switching losses if the frequency is high.

Looking forward to hearing how your fun project progresses!

7
For the voltage waveform, I'd guess it's distorted by inductive voltage drop in the ground.  If you can ground the scope probe at the IGBT brick and measure the bridge output voltage near by on the same brick, you'll likely see less distortion.

My first guess for low performance is mismatched primary-to-secondary frequency.  But that's entirely a guess.

Hopefully you'll get more responses from others here who have built more similar coils.

8
I haven't personally built any antenna-feedback systems, only current-sense feedback.  However, I'd suggest adding a high-value (~1meg) resistor from input to output of the antenna inverter (pin 1 to 2 of U2 in the original schematic).  The antenna input node (U2-1) has no defined DC bias level, so may be anywhere from 0 to 5V depending on diode leakage currents.  The first half-cycle of drive has to generate enough voltage to switch the inverter from any initial voltage point.  The feedback resistor will keep the DC bias near switching point.  With the HC14, the feedback resistor will cause it to oscillate even with no feedback signal.  The oscillation frequency can be set to something near your intended 220kHz by adjusting the feedback resistor value.  (I can't calculate the value ahead of time because it depends on total capacitance of the HC14 and D7 and D8, and on the hysteresis of your particular HC14 chip.)  With the self-oscillation frequency anywhere close to 220kHz, it can build up much more voltage in the top load before the antenna feedback is enough to synchronize the oscillation.

9
Spark gap Tesla coils / Re: SGTC MK1 - An Accomplishment in Progress
« on: November 19, 2019, 05:47:34 AM »
Are you using standard TV flyback transformers as I've been assuming?  Those almost always have internal diodes, so output DC, positive on the HV lead and negative on the return pin.  If you were using bare AC flyback transformers, then you would have HV diode(s) between the secondaries and spark gap.

The normal DC flyback transforms wouldn't work back-to-back in series.  Both ends would be positive, so no voltage between them.  Your series schematic doesn't show diodes either internal or external.  Is that what you are simulating - no diodes?  If the simulation values (inductances, capacitances, etc.) are reasonable, you won't be getting much voltage on the MMC, since it can charge for only one half-cycle of the flyback frequency.

10
The 5V supply current is so low, perhaps 1-2mA, that it shouldn't make any difference where the 7805 input is wired.
Yes, 0.1 in parallel with C9, C10, and C11 is a good idea.

The TI data sheet for the drivers suggests 0.1uF for the input side (pin 1 to 4) and 1.0uF for the output side (pin 5 to 8).  Section 10 of the data sheet shows suggested ECB layout, showing that input circuitry should be referenced to pin 4 and output load (gate drive) referenced to pin 5 ground.  If you make a 1-layer ECB, I'd suggest a combination of ECB and proto-board style hand wiring.  Use the ECB layer primarily for ground plane, especially under and around the driver chips.  Make the signal connections with wires, laid flat against the ground plane.  (I haven't looked recently, but there used to be vectorboard available with a ground "plane" included - actually a 0.1" square grid of ground wires running between the 0.1" grid of holes.  That's another option rather than making a 1-layer ECB.)  I'd still recommend two bypass capacitors per driver chip, soldered directly from the power pins to the corresponding ground pins or to the ground plane.

A side note:  Ceramic capacitors are generally best for high-frequency bypassing.  However, spec's can be misleading.  A "1.0uF" capacitor rated for 25V may drop to 0.5 or even 0.3uF at 12V, and down around 0.1uF at rated 25uF.  A few manufacturers are starting to publish at least typical data for this voltage degradation of capacitance.

Concerning the size of the local driver pin 5-8 bypass capacitor, it depends on the wiring inductance back to the 12V bulk capacitor C10.  I'd suggest at least the 1.0uF that TI recommends, especially for a hand-wired board that will have more wiring inductance.

Driver heating is the trickiest problem to figure out.  If it's caused by high-frequency oscillation, then adding tightly-coupled bypass capacitors and separating the input-side circuitry from the output-side should stop the oscillation and fix the heating issue.  (Also need a good low-inductance connection between the two driver chips output grounds - pins 5, since the gate-drive current flows from one chip to the other, so needs a low-inductance return path.)

If the heating is just due to normal gate-drive power, then, yes, it will get worse when C6 is increased to 1-2uF.  For that reason, I'd start even lower than 30% duty cycle 10% or less.  You could run some initial gate-drive heat testing without the IGBT power connected.  (That will change gate-drive overshoot some, but gate-drive power should still be 80-90% of it's full-load condition with 170V on the IGBTs.)  Without IGBT power, it will be safer to measure 12V current, waveforms at the driver chip outputs, etc.

The TI UCC27321/2 data sheet lacks information on drive current vs. voltage during the transitions, making it difficult to calculate how the gate power will split between the gate resistors and driver chip.  With 47-ohm gate resistors, the drivers should see ~20-30% of the power.  With 10-ohm gate resistors, it will be higher, perhaps 40-50% of the power in the driver chips.

To make calculations even harder, this data sheet appears to have some self-inconsistencies.  Max die temperature is listed as 150C.  Thermal resistance to ambient is listed as 55.9 C/W for the PDIP package.  At 2 watts, that's 112C rise, which would allow operation at 38C ambient.  However, the power dissipation table lists maximum of 350mW at 25C ambient, roughly 1/6th of what the thermal resistance value would imply.  If the correct answer is 2W, you'll probably be fine once any oscillations are fixed.  If the correct answer is 350mW, then these drivers aren't sufficient for your application except at low duty cycle.  Anyone else see how to reconcile these data-sheet values?  Am I making an obvious mistake here?

BTW, there's another apparent mismatch in the data sheet:  Figures 7 and 8 show rise and fall times vs. supply voltage.  Figures 9 and 10 show rise and fall times vs. load capacitance.  For the 10nF load condition of the first two graphs, the ~20ns rise and fall times are not close to the 10nF (right edge) of the other two graphs, which show ~27ns rise and ~170ns fall.

Your diligence on this student project is impressive!  This is good experience for the engineering world, where projects are always more complex than they initially seem.

11
Spark gap Tesla coils / Re: SGTC MK1 - An Accomplishment in Progress
« on: November 18, 2019, 01:59:24 AM »
Your proposed series connection could work, but it requires the upper secondary winding of your custom transformer to handle 15~30kV relative to the lower secondary and primaries, which are essentially at ground potential (compared to 30kV).  15~30kV is whatever voltage a single flyback output can generate.  If you can construct a custom transformer with such HV insulation, I'd suggest making your own flyback large enough to drive your project directly with one stage.  If you do try your series design, I'd add a connection from the negative (return) secondary pin of the upper flyback to it's primary.  This provides a path for any leakage current from you custom input transformer, preventing it from damaging the upper flyback.

Starting with a single flyback sounds like a good idea.  If you get enough voltage to trigger your spark gap, then series isn't needed.  Add a second in parallel if you want to double the firing frequency.  BTW, do you know roughly what voltage your spark gap is set for?

Small fans are quite cheap these days.  I'd suggest placing one blowing directly on the flyback(s).  Of course, there's still higher temperature inside than on the surface, but a fan will significantly improve the power capability.

12
Since your gate waveform looks fine w/o IGBTs connected, I believe your gate-drive transformer core is fine.  You could run it for a while unloaded (no IGBTs) to make sure the core doesn't warm up, which would happen if it has too much eddy-current or hysteresis losses for 220kHz.  With C6 at only 0.1uF, the unloaded waveform would be obviously distorted if inductance were too low.

Just realized, 0.1uF is not enough for driving your four IGBTs.  Thier total gate capacitance is ~66nF (0.066uF).  That explains almost all of the gate-voltage drop when loaded (when IGBTs are connected).  It's forming a capacitive divider of 0.1uF and 0.066uF.  To avoid much voltage drop, C6 should be much larger than the total gate capacitance, at least 1uF in your case.  2.2uF would be better.

Wiring inductance in the gate circuit effectively adds to leakage inductance, so should be minimized as Mads suggested.  Wiring inductance is almost always parasitic, so best to minimize everywhere.  (Although not a critical issue at this point, even the 0.8uH gate transformer leakage inductance will limit gate switching time to ~400ns.  A common technique for lowering leakage inductance is to wind with four twisted pairs, 8 wires total.  Cat5 or similar cable is easy for this.  One half of each twisted pair is used for gate-connections.  The remain four wires, one from each pair, are connected in parallel and used for the primary.  It's similar to what you have, except the primary has four wires instead of one, and each primary wire is tightly coupled to one secondary wire.)

For circuit board layout, the best option is to reserve one layer for a ground plane, with as few gaps as possible if any are needed to make short connections bridging traces on other layer(s).  That ground plane will help some with driver heat-sinking too.  (There are driver chips in packages designed for heat-sink attachment, but hopefully that isn't necessary for your design.  That's presuming the power issue is resolved by eliminating oscillation or whatever the issue turns out to be.)  I'd also suggest adding more bypass capacitors.  Duplicate C4 and C5.  Place one across the input side, pins 1 to 4.  Place the other across the output side, pins 5 to 8.  Place caps as close to the chip as possible.

Low ground inductance is good for breadboards too.  If possible, copper tape strips for ground.  Good bypassing is also important.  I often solder bypass capacitors directly to the chips, especially ones with high switching current spikes, including the drivers and 555.  As a first experiment, I'd try soldering bypass capacitors to your driver chips, 2 per chip as mentioned above.  You could leave C4 and C5 in place, and just add more bypass caps directly at the chip pins.  They can be soldered to the pins under the breadboard, or directly to the sides of the chips on top (assuming you're using the DIP packages).  Just adding those caps might be enough to fix oscillation if that is the reason for excess driver heating.

Another suggestion for breadboarding this circuit in particular:  The two driver chips should have Kelvin ground connections much as the IGBT emitters.  Hopefully the two driver chips are reasonably close.  Wire the two chip's input-side ground pins (pin 4) together.  Separately wire the two output-side ground pins (pin 5) together.  Connect the two ground wires together at one location.  Connect the input circuitry only to the input-side pin-4 ground wire.  You could do the same with the +12V power, but that's much less critical.  The logic threshold voltage is much closer to ground than to +12V.

One crude method to look for oscillation or rings that are too fast for your scope:  Place a small capacitor, 100-1000pF, at the probe tip, from tip to ground.  Add a fast signal diode in series with the scope tip.  Probe with the open end of the signal diode.  Switch the diode direction to probe for peak positive or peak negative voltage.  If the gate-driver outputs get much above +12V or below ground, then there is an issue with fast oscillation or ringing.

13
High gate resistors does lead to slow switching, which leads to more heat in the IGBTs.  The higher gate resistors should reduce gate-driver chip power, moving that power to the gate resistors themselves.  So, I'm guessing there's some issue with the driver circuitry itself.

Can you post images of the driver board, or the board layout?  Parasitic inductance on board traces, especially ground pins, may lead to local high-frequency oscillations.  Such oscillations, which could be 30-300MHz, may explain both problems, hot driver chips and gate drive voltage dropping to +-7V.  Depending on the scope and probes, the oscillation may be too high frequency to show up.  They'll also be filtered out by the gate resistors.  However, the resulting gate waveform will not reach 12V because it is the average of the high-frequency waveform.  (I've experienced that issue in several of my earlier hobby circuits, though they weren't Tesla coils.)

Not related to the above speculation of high-frequency local driver-board oscillations, but here's a bit more on IGBT emitter connections:  Dual contacts to the emitter are referred to as "Kelvin connections" in case you want to research that more.  To be clear, I've updated my half-bridge circuit sketch to show that the diodes across the IGBTs are part of the high-current connections.  The gate-drive return shouldn't share any wiring with any of the high-current paths:


14
UF4007 is a nice fast diode, but rated for only 1A average, 30A non-repetitive peak.  The diodes don't see as much average current as the IGBTs do, but they do see almost the same peak current, at the end of each enable period.  If the drive timing isn't phased close to the zero-current point, then the diodes can see quite a bit of average current too.  So, I'd suggest multiple UF4007 diodes in parallel for each IGBT.  The original instructable used 8A diodes, so 8 in parallel might be a good starting point.

The diodes should be physically close to each other (and to the IGBT) with good thermal contact to keep the diode temperatures matched.  Diodes generally have a negative forward-voltage vs. temperature curve, so the hot diode will hog current and get hotter.  Most heat is conducted through the leads, so solder the leads all together or to a short piece of heavy copper wire or plane such as a piece of 12AWG wire.  (One wire piece for cathodes and another for anodes.)

As with the diodes, most of the driver chip heat is conducted out the leads, especially the negative power lead (usually - not certain about these chips).  That makes it hard to add more heat dissipation capability without changing the circuit board design to have larger copper areas for those leads.  You can gain a bit of cooling by gluing a piece of copper or aluminum to the top of the packages.  A small fan blowing directly at the chips will help too.

Scoping the waveforms on the driver chips, especially the outputs, would be a good idea too.  Perhaps some ringing is causing more output switching than the intended 220kHz.  Especially with the voltage dropping to 7V, the power shouldn't be all that high.  (That's presuming the 12V supply is dropping as the cause for lower gate voltage.)

I've seen gate drive resistors on either side, but by far most often on the secondaries of the gate drive transformer, one set per IGBT.  Often the resistors include a diode and second resistor to make turn-off faster than turn-on:


Other members here likely have more experience with gate-drive transformer/resistor combinations than I do.  Hopefully you'll get more replies if my suggestion isn't optimum.

Another thought on the driver heating:  Each IGBT emitter has two connections, one for the high-current path (to Vbus or MMC or Tesla primary winding), and another for gate-drive transformer.  These two connections should separately wire to each IGBT emitter pin, as close to the IGBT case as possible.  Larger IGBTs usually have two emitter connections, one for the high-current path and another just for gate-drive return.  If the high-current path shares wiring with gate-drive return, then the inductive voltage drop due to the high-current well be added to the gate waveform.  I've tried to show that in the above gate-drive schematic, with the emitter connection splitting just under each IGBT.

One final recommendation:  Until everything is working properly, run well under 50% duty cycle, say 5%.  Mistakes are less likely to fry parts when they are running cooler.

15
Spark gap Tesla coils / Re: SGTC MK1 - An Accomplishment in Progress
« on: November 16, 2019, 04:13:35 AM »
Placing two flyback transformer secondaries in series will almost-certainly fry one of them.  Flybacks are generally designed to have the HV negative (return) terminal somewhere near the same potential as the primary.  In series, one of the secondary negative pins will be many kV away from it's primary.  Internal insulation on the negative side isn't designed to handle that voltage.

Paralleling both the primaries and secondaries should work, and is probably closer to what you need.  The output voltage is likely enough from one flyback.  Using two in parallel will double the output current, so charge your MMC in half the time.

Yes, perf-board will work fine.  The string method is faster for the large quantities I needed.  Yes, the inductors do interact magnetically with adjacent ones.  Alternating the orientation (180 degree rotation) for radial-lead inductors as I used increases inductance (makes magnetic loops).  That's what I wanted.  It does lower saturation current, however.  Making a string of radial-lead inductors all in the same orientation lowers inductance, but increases saturation current.  I'd recommend making the entire string either the same or alternating, rather than mixing the two options.  If the inductors are spaced out a few mm, then direction won't matter much.  The inductors I used have a small white dot mark on top to show orientation.

For the inductors I suggested previously, resistance isn't enough to make much difference.  For example, the 150-ohm ones, 100 in series will have 15k ohms.  If charging at 30mA from the flyback(s), that's 450V drop, not too much compared to ~30kV.

To avoid excess voltage across any given inductor during the spark discharge, stray capacitance within the inductor string should be reasonably uniform.  I'd suggest a physical layout on your proto-board matching this schematic:


In other words, don't wire a zig-zag.  Make series-connected rows of inductors, wiring the right edge of each row to the left edge of the next row.  That way the electric field will be roughly-uniform from top-to-bottom.  Leave a few mm between rows, whether or not you decide to space out inductors within each row.  Mount the proto-board inductor string in a plastic case or otherwise spaced away from metal.

Good luck with your fun project!

16
Hopefully you are including D1 and D2 from the schematic, as those old IGBT parts don't have an internal diode.  (Much faster IGBTs with internal diodes are available, but I presume the price was right for the IXGH40N60A parts.)

As with most IGBTs, the IXGH40N60A parts are designed for 15Vge.  That's where the current ratings are specified.  12V is a bit low, although probably OK for typical parts.  Most silicon FETs are designed for 10Vgs.  The instructable uses FETs, so 12Vgs is plenty including margin.

The gate charge for one IXGH40N60A is 200nC for 0 to 12V gate.  Although not specified, it's likely another 200nC for the -12Vge to 0Vge half of the gate-drive waveform.  So 400nC total per IGBT, or 1.6uC for all four parts of the bridge.  That charge is required on every edge of the drive waveform, so 440kHz repeat rate.  440kHz * 1.6uC = 704mA.  704mA * 12V = 8.45W, shared between the two gate drivers and the four gate resistors.  Not too surprising that drivers run warm.  This is based on continuous operation.  Many coils are pulsed (enabled and disabled) to play tunes or whatever.  The gate drive power will drop by the duty cycle of the enable pulsing.

My only guess for your +-12V to +-7V issue is that the 12Vdc supply isn't holding up to the 700+mA cuyrrent.  Have you tried measuring the 12Vdc supply while running?

Did the scope probe showing the ring have a missing ground connection?  I'd be a bit careful of assigning ringing to probes.  Gate-drive-transformer leakage inductance and general wiring inductance can cause such ringing.  For wiring inductance, the ringing may be present at one end of a cable and not at the other.

17
Dual Resonant Solid State Tesla coils / Re: Problems with my first DRSSTC
« on: November 14, 2019, 05:35:19 AM »
The input bridge current depends on how hard you are going to run your coil - frequency and width of the enable pulses.  If you are measuring line current, just keep it under 8A or buy higher current diode bridge.  BTW, I recently bought some cheap bridges from China on EBay, supposedly 50A at 1000V, but they were worthless, perhaps good for 5A.  Years ago I'd gotten some of these that worked OK, at least at 15A.

Once the driver is swapped to non-inverting, it will require reversing either the CT feedback or GDT terminals to get the feedback phase back to correct.

18
Spark gap Tesla coils / Re: SGTC MK1 - An Accomplishment in Progress
« on: November 14, 2019, 05:25:57 AM »
Grounding the flyback HV return pin might help slightly.  At least it should prevent any internal arcing from the flyback secondary back to primary.

A string of small inductors isn't all that hard or expensive.  I needed 3000+ because I needed 24 inductors each capable of 48kV.  You need only one inductor at perhaps 30kV or whatever your spark-gap firing voltage is.  100 inductors is probably fine.  Each inductor sees only 300V, which shouldn't be enough to break down even the thin magnet wire insulation.  200 inductors would give lots of margin.  Here's links to a couple Digikey pages for inductor parts that should work, roughly $20 and $30 for 100 parts:
https://www.digikey.com/product-detail/en/taiyo-yuden/LHL08TB153J/587-5891-1-ND/7675011
https://www.digikey.com/product-detail/en/bourns-inc/RLB1014-104KL/RLB1014-104KL-ND/2561370
Checking more distributors might turn up a lower price for these or other similar parts.  I just searched for the lowest cost and highest inductance that could handle 40mA.  Do you know what your flyback output current is?  I was just taking a guess that it wouldn't be above 40mA.

The key detail to making the inductor string is to have uniformly distributed stray capacitance.  That keeps the voltage evenly distributed.  I made a soldering fixture of a string of 6.35mm magnets.  Cut the inductor leads to ~6mm, then lined up a row on each side of the magnet string, staggered so the leads touched in a series configuration.  Then it was easy to run down the string bonding the touching lead pairs with solder.  I made 19-long inductor strings this way, 10 on one side and 9 on the other side.  Each string went into 1" heat-shrink tubing.  The resulting insulated strings were layered (stacked).  I'd share pictures, but my fixture is buried in my storage shed at the moment.  If you want to go this route, I'll get it out and add further description. 

19
I used a 50 microamp analog DC current meter.  I made a full-wave-bridge of 4 1N4007 diodes into the DC current meter to rectify the AC to DC.   Then I made about a 600 megaohm resistor by putting 33 20 megaohm resistors in series and place this in series with the full wave bridge.  To prevent end-to-end arcing breakdown between the resistors, I laid them out in an undulating pattern and potted them all in epoxy.  This was able to read up to 30 kV peak, but you could add more resistance for a higher range.

If I understand your circuit, it's measuring average rather than peak.  With the diode choice and stray capacitance of the resistance string, it would work well at line frequency, but not at typical flyback frequencies.

20
I haven't personally tested above 120VAC/170VDC.  A friend in Germany prototyped a ZVS induction heater running on 230VAC (325V peak) using 1200V SiC FETs.  He's using my older circuit topology with small low-voltage PFETs to reduce gate-drive power and allow stiffer pull-up resistors (shorter turn-on time).  He's planning to make a commercial product, so likely doesn't want details shared here.

Even 400VAC (566V peak) should be possible using 2500V IGBTs.  It would likely require gate-driver ICs and more complex control.  I think of simplicity as the key advantage of ZVS Royer oscillators.  Once the control gets more detailed, is there a reason for ZVS over bridge ZCS?

profdc9:   Is there a bridge ZVS topology?  I'd love to learn about that.  Or, did you mean ZCS for the bridge?

BTW, 1200V FETs at 325V peak input has very little headroom.  Roughly the same ratio as my 170V input here using 600V IGBTs.  Success with such tight margins required two design features:  Lots of TVS protection and controlled oscillation startup.  The latter is the reason for my low-power idle oscillation before the main switch is closed.

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