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Messages - davekni

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1
Phase lead looks great now!  Double-check that when you get back to full current, as phase lead may be current-dependent given the driver's clamped feedback load.

If I recall correctly, you are using 10 ohms on your scope current transformer.  The driver's over-current looks more like 1 ohm plus two diode drops.  Perhaps that's the scaling issue - a different driver load resistance than what you are using in calculations.

2
Electronic Circuits / Re: Power MOSFETs with fast recovery body diodes
« on: April 01, 2020, 08:47:11 PM »
Usually there's a trade-off that faster diodes have higher forward voltage drop.  To my surprise, that isn't the case for these parts.  I did notice that the avalanche energy rating was a little lower for the fast-diode FET - no idea why.

3
Electronic Circuits / Re: TVS diode selection for 400v transistor
« on: April 01, 2020, 08:24:06 PM »
Do you have any MJE13007 parts you haven't used yet?  You could try it at -5V instead of -9Vbe.  The beta degradation is tied to reverse BE leakage current, which typically rises sharply as you approach the -9Vbe spec limit.  I don't know if switching speeds or any other parameters degrade with beta.  You might find that an MJE13007 part operated within recommended conditions works well.

Yes, MJE13007 is an unusual part relative to my experience.

4
Electronic Circuits / Re: Pt100 oddities
« on: April 01, 2020, 08:16:59 PM »
If I'm understanding correctly, you are down to calibrating the cheap Pt100 sensors, as your electronics are repeatable.  If so, a scale factor is more appropriate than an offset.  Being cheap, these are likely thin-film RTDs, perhaps rejects where the deposited platinum was too thin so couldn't be laser-trimmed to 100 ohms at 0C.  Measure the actual resistance at 0C.  Calculate the required sale factor to get to 100 ohms.  Then multiply all resistance readings by that scale factor prior to converting to temperature.

I think platinum resistance/temperature curve extrapolates to 0 ohms at -260C.  So, if your 4-20mA results are already in degree C units, add 260, multiply by the scale factor, then subtract 260.

5
Are the sine-wave traces of the primary voltage, the junction between the MMC and primary coil?  Presuming so, it looks like a bit too much phase lead, although that's definitely better than phase lag.  Ideally the H-Bridge voltage steps on top of the primary voltage would be just 5-10 degrees or so before the peak.  Looks like perhaps 25 degrees now.  Would need to zoom in closer to tell.

Yes, the low voltage makes IGBT capacitance higher, so ringing tends to be worse.  Lower current may change the behavior of your phase-lead circuit, since the driver board's input impedance isn't linear with voltage.

6
Beginners / Re: A few (stupidly basic) questions
« on: April 01, 2020, 01:32:54 AM »
Mads,

I'm curious about your line: "I have only built mains powered SSTCs as I find the low voltage versions to be harder to get to work and they also seem to have lower spark-length-efficiency, I am a sucker for long sparks".  If the power levels are equivalent and the part impedances are all scaled appropriately (larger capacitors, lower inductances, etc.), what makes the difference?  The best guess I'd have is that wiring inductance through the primary circuit (H-Bridge to primary coil) isn't reduced enough.  Low impedance would require twisted pairs and tight layout.  Do you have other possible reasons?

Thank you.

7
Thank you for the scope traces!  Do you know if both gate-drive outputs of your driver board always switch together?  I'm guessing so from the traces, but it's a bit hard to tell.  Pausing only one half of the H-Bridge makes a more efficient pulse-skip mode.  It makes the H-Bridge output go to ~0, rather than to inverted, so makes the current ramp down more slowly.  Your scope traces show current ramping down fairly rapidly during pulse-skip periods, leading me to guess that the two gate-drive outputs are the same.

There are two possible issues with this form of pulse-skipping.  First is the same bulk-cap to H-Bridge wiring inductance we've discussed already.  It's now being driven multiple times in a row.  If the skip pattern happens to hit the resonance of that inductance and your local H-Bridge snubber capacitance, the local VBus voltage peaks at the bridge will get higher.  However, the one scope trace of an H-Bridge output doesn't show evidence of such a problem.  What current was running for the H-Bridge output scope trace?  It may be worth scoping an H-Bridge output more as you set current limit higher again.

The other possible issue with pulse-skipping is the positive ring on the gate-drive signal when transitioning from negative to 0V.  (For example, see the upper trace of the scope plot labeled "Left side gate drive and right side gate drive".)  If that spike is enough to momentarily turn on an IGBT, before the opposing one is off, it would cause current shoot-through.  (The IGBT turning off will do so more slowly than normal, as it's gate voltage is going from positive to 0, not to -18V.)

Yes, there does appear to be an issue with current limit scaling.  That could be the entire issue.

Phasing appears to be about at 0 degrees.  It is best to have a little bit of phase lead, so IGBTs turn off slightly before current reaches 0, allowing the remaining bit of current to cause the voltage swing before the opposing IGBTs turn on.  So, yes, slightly more phase lead would be ideal.

8
Electronic Circuits / Re: TVS diode selection for 400v transistor
« on: March 31, 2020, 04:44:05 AM »
The MJE13007 data sheet has a graph called "Maximum Reverse Bias Switching Safe Operating Area", figure 7.  It shows the allowed current at turn-off vs. Vce, with different graphs for different levels of Vbe, 0V, -2V, and -5V.  I see that it's better with more negative Vbe.  Hadn't ran into that situation before.  It may improve farther to -9Vbe, but it's also possible that the curve would reverse direction and allow lower Vce at -9Vbe.  If the Vce punch-through voltage is only slightly above the 700V avalanche-breakdown voltage for Vces, then the more negative base could bring the punch-through breakdown voltage below 700V.  So, -5Vbe is good, and -9Vbe MIGHT be better.  (Punch-through is when the reverse-biased depletion regions for C-B and B-E meet in the middle of the base region.  It's a different voltage breakdown mechanism than the more common avalanche breakdown.)

Notice that -9Vbe is the maximum allowed.  An 8.2V nominal zener plus diode forward drop will be typically about 9V.  Worst case it could be a bit higher.  For a one-up hobby project, that's likely fine.  Might drop to a 7.5V zener if making more than one.  One other consideration from my brother Dan:  Many bipolar transistors have their properties, particularly beta, degrade when operated with reverse Vbe for extended periods.  This is true especially when Vbe is close to breakdown voltage.  Dan's experience is with high-frequency BJTs, not power devices, so this may or may not be significant for this device.

That zener looks plenty capable for several hundred mA.  No need to use a TVS instead.

9
A documented driver is a good idea.  If you want to continue learning about this driver, while you have over-current set low, scope the gate-drive outputs of your driver.  It may be tricky to get all the information with a two-channel scope.  Can you get into a stable mode, perhaps w/o secondary, where the pulse-skipping is consistent from one firing to the next?  It would be interesting to see how this driver runs pulse-skip.

10
Electronic Circuits / Re: TVS diode selection for 400v transistor
« on: March 30, 2020, 06:35:13 PM »
Yes, Vces (700V) just needs the base pulled down to the emitter voltage to sink the collector-to-base leakage current.  I'm rusty on my semiconductor physics, but having the base below emitter voltage by almost Vbe breakdown might actually make things worse than 0Vbe.  I need to consult my brother, as he is more versed in such matters.  I'd suggest clamping the base voltage to one diode-drop below emitter just to be safe.  There's no value in going farther negative.

The Vces is typically a static (DC) test.  There may also be issues during turn-off, when there's still some minority carriers in the base region (base storage charge).  Again, I'll need to think about this more and/or consult my brother.

11
Beginners / Re: A few (stupidly basic) questions
« on: March 30, 2020, 02:23:23 AM »
Curtis,

Line voltage can be dangerous, but microwave oven transformers (MOTs) are MUCH more dangerous.  A man died here in Wilsonville a few years ago using an MOT for burning lichtenberg figures, accidentally touching the wires.  (I made an SRSGTC using two series-connected MOTs, 2.5kW average power.  That's my toy that scares me the most to run.)

Yes, SSTCs are generally safer.  And, yes, you can run one easily from 60V 30A - that's plenty of power.  Just requires higher current FETs or IGBTs, larger DC blocking capacitor, and fewer turns on the primary.

Can't comment much on RF regulations, which generally start at 30MHz for radiated, lower for conducted down line cord.  I suspect most Tesla Coils violate such regulations, at least when making strikes to a ground target.

If running with an interrupter, so pulsed operation, it's quite possible to have higher peak currents while keeping the average under 15A.

If you are using a bunch of scavenged parts (which is great), the first step is to measure them.

12
Mads, thank you for the compliment, and especially thank you for moderating this forum!  I was thrilled to come across it last September.

"So it looks like the exact 5 IGBTs survive each time. It's always the 4 IGBTs on the left and the one on the top right. All the other 3 fail. I think this might be because I'm using 550V TVS, and maybe the IGBTs on the left got "lucky" and got the ones that triggered on lower voltage. I should probably switch to 440V. However, nothing bad seems to have happened yesterday, and I tested it at higher voltage. Maybe the extra current contributes to higher voltage spikes?"

Yes, most spike voltages are roughly proportional to current.  IGBT capacitance is voltage-dependent, and other IGBT parameters depend on voltage and current, so it's not completely simple.  In particular, the low-frequency (wide) voltage spike at the end of the enable pulse caused by bulk-capacitor to bridge wiring inductance is quite linearly proportional to current.  That would be my first guess for this latest frying.  Uneven current sharing is also a possible cause as Mads pointed out.

Yes, reducing the TVS voltage would be a good idea.  If the voltage spike has too much energy for a lower-voltage TVS to survive, at least a fried TVS diode would provide information that there was an over-voltage event.  (Although it's also possible that an over-voltage event is secondary, caused by the first IGBT frying, before the opposing one(s) fries.)

13
Electronic Circuits / Re: TVS diode selection for 400v transistor
« on: March 29, 2020, 09:08:39 PM »
The 258V clamp voltage will increase if the diode is required to clamp repetitively, enough to get it hot.  But that is a great ratio from 250V rating to 258V clamp even for single-pulse, a good low-impedance part.

Yes, I'd go for 300V.  It isn't quite guaranteed worst-case (414V), but likely to be fine.

The best I can tell, the 1500W diodes are equivalent to 2.5 paralleled 600W diodes.

The only real down-side to bidirectional TVS diodes that I'm aware of is slower clamping in the ns range due to forward recovery time of the forward-biased diode.  Capacitance is generally lower for bidirectional TVS diodes due to being two in series.  The clamping speed should be no issue for your use, since the flyback capacitor limits voltage slew rate already.

14
Did you turn off pulse-skipping mode?  Or is that not possible?  At half inductance and the same frequency, you'll be running about twice the current, so much more likely to high current limit and pulse-skipping mode, which may fry more IGBTs.

15
For grounding, I'll emphasize a line from my last answer: "  (Of course, for the bridge, this works only when using a bench supply or transformer, not when powered directly from line voltage.  Once running on line voltage, don't ground the negative side of C12, and don't clip the scope probe ground to anything on the bridge circuitry.  Leave the driver grounded even with direct line voltage on the bridge.)"

As long as you are running from your hy3005 supply or a transformer, ground the negative side of C12, and clip the scope probe ground to that node as well.  The ground terminal of your hy3005 supply is a good place to ground the bridge.  Don't do that once running directly from line power!  For scoping any circuit, it's best to have the scope probe ground clip connected to the ground node of the circuit you are probing.  If the circuit doesn't have a ground node, such as the h-bridge when connected to line voltage, then a differential probe is needed.  (Floating the scope ground can work instead, but only when the probe's ground lead is connected to a low-impedance circuit node, such as the negative side of C12, where injected noise won't be a problem.  Floating scope ground will now have rectified line voltage, so is a shock hazard.  And the other scope probe's ground clip will be at the same line voltage.  So, don't scope any other signals at the same time - only one signal when using floating ground.  In general, floating scope ground is risky, so I recommend not floating your scope ground.)

Yes, differential probes are useful for scoping many things, such as speaker terminals as you mentioned.  Old amplifiers usually had one speaker terminal at ground and drove the other with the audio signal.  Many amplifiers today drive both sides of the speaker with opposite-polarity audio signals.  So you shouldn't connect either speaker terminal to the scope ground.  For most audio amplifiers, the speaker signal on either side is sufficient for probing, because the other side will be an inverted version of the same waveform.  If you need to scope across the speaker, either use a differential probe, or use both scope probes and the waveform-subtract math function of the scope.  That does the same thing as a differential probe, although not quite as accurately as a good differential probe.

The H-Bridge output should be a square wave with a frequency of 240kHz (your resonant frequency).  It's always connected to either the negative side of C12 or to the positive side of C12.  The phase information I'm interested in is when the square wave edges are timed relative to the secondary high-voltage sine wave.

When the bridge is powered directly from line voltage, then there will be some rectified 50Hz sine wave signal on top of the much higher frequency square wave signal.  Since the frequencies are so different, scoping the bridge output can be useful even when line-powered.  (The scope probe ground needs to be left unconnected when probing a line-powered bridge!)

Sounds like the high-frequency signals of your SSTC are confusing the power supply circuitry.  Connecting the supply negative output to the supply ground terminal may fix that issue, making the supply behave correctly.

If I missed answering anything, please ask again.

16
Yes, the radio station thought was unlikely for you.  An AM station antenna would be large, so obvious to see.  It was just childhood memories for me.

As long as it's only the "ground" nodes being grounded, not literally "everything", then nothing will blow up.  The scope should be grounded, the driver ground (pins 4 and 5 of the UCC chips etc.) should be grounded, and the bridge negative supply (negative side of C12) should be grounded.  Then the scope probe ground can be clipped to either the driver or bridge grounds for measuring voltages.  You cannot connect the scope across two non-grounded nodes, such as measuring the voltage across C4 of the driver, because that would be grounding one side of C4.  Always measure voltages relative to ground, with the scope probe ground clip on one of the ground nodes.  (Of course, for the bridge, this works only when using a bench supply or transformer, not when powered directly from line voltage.  Once running on line voltage, don't ground the negative side of C12, and don't clip the scope probe ground to anything on the bridge circuitry.  Leave the driver grounded even with direct line voltage on the bridge.)

It sounds to me like your power supply has an internal failure causing the 44V jump.  Might still be usable though.  If its current limit still functions, that would add a bit of protection over using a transformer directly, even if the voltage isn't fully adjustable.  The supply's internal failure might make current limit non-functional as well.

Yes, higher coupling is generally good for an SSTC, until the point where the primary gets so close to the secondary that voltage starts arcs across the gap.  Look in the dark at the bottom of the secondary and at the primary to see if there's significant corona discharge.  That indicates you are close to a problem there.

As you've found, low coupling doesn't generate enough secondary voltage to couple into the antenna and start oscillation (or lock oscillation to the resonant frequency in this case with self-oscillation).

For phase measurement, here's what I was asking in reply 24:  "Coarse phasing (180 degrees or not) is easiest by trying both ways to see which one locks frequency.  Once the coarse phasing is correct and the coil is running with feedback, I suggest measuring the more subtle phase shift.  Scope the H-Bridge output with one probe and use the other probe as an antenna - just hanging in the air somewhere around the coil.  Ideally the H-Bridge output switches at or just before the peaks of the top-load voltage (which the floating scope probe is picking up).  Leave the floating probe separate from the feedback antenna to avoid changing behavior."

Now that you are running, phase measurement isn't critical.  I am personally interested in the phase measurement, however, not having any experience myself with antenna feedback.  (Antennas seem to likely to pick up other stray signals, so I've always used current feedback.)  Thank you for your willingness to provide the traces!

17
That's lots of symptoms.  I certainly won't be able to tell remotely exactly what happened.

"Floating" scopes or other equipment aren't completely floating.  There's always capacitance to the line neutral and hot wires in the scope's power supply.  The scope is likely injecting more noise when floating.  I recommend keeping your scope grounded, and keeping the driver circuit grounded.  During this initial bring-up with a bench power supply, ground the bridge negative supply as well.  With ungrounded circuitry, part of the signal the antenna sees is the noise on its local "ground" reference, the negative driver supply terminal.

(Do you happen to live close to a commercial radio broadcaster?  As a kid we lived about 1km from an AM radio station.  Any sort of antenna picked up obvious amounts of that ~1MHz signal.)

Especially without the secondary in place, the antenna may be picking up noise from any source, including floating "grounds".  The most problematic may be if the gate wiring is long enough and/or close enough to the antenna that gate-drive becomes the feedback.  There's nothing in that circuit to prevent oscillation at high frequency, which would likely happen if picking up feedback from the gate drive wires.  If you ground everything, at least that eliminates some of the noise sources, so it would be easier to see any high-frequency oscillation.

During normal operation, the high secondary voltage dominates over any other noise sources the antenna may pick up.  Testing without the secondary can still be possible as long as things are grounded and the gate-drive wiring is shielded from the antenna.

Concerning 44V, that may be a digital meter getting confused by high-frequency noise.  Cheap meters have little shielding internally.  Usually I see random quickly-changing values in such circumstances rather than voltage errors, but I have occasionally seen just an offset as you are seeing.

18
Electronic Circuits / Re: TVS diode selection for 400v transistor
« on: March 28, 2020, 04:44:17 AM »
Yes, worst-case specifications for TVS diodes don't allow tight clamping.  I've purchased quite a range of TVS diodes, finding them handy to have for my personal stock.  All have been quite close to their nominal voltage at low current and room temperature.  So, a P6KE300 is likely to have a breakdown right around 300V.  However, that voltage will depend on current, and especially on temperature.  The clamping voltage and current is specified with a total energy that heats the diode to its maximum 150C or 175C.  As with most silicon avalanche-breakdown voltages, it goes up with temperature.  So, the P6KE300's worst-case clamping voltage of 414V might not quite protect your 400V FET.  That depends heavily on the TVS current pulse duration and repeat frequency - in other words how hot the TVS diode gets.  Also, if your FET is running hot, it's breakdown voltage will be above 400V.  And, some FETs have some avalanche energy capability.  Overall, P6KE300 is probably the best choice.  Definitely not P6KE350.  Use the P6KE250 if you want to be extra safe with clamping voltage.  Still, be careful about letting the TVS do too much clamping, keeping it under its average power dissipation rating.  Or, go to 1.5KE300 for more power and higher peak current capability.

19
Are your fast diodes on heat sinks along with the IGBTs?  If not, are you sure the diodes are all still good?  I'm wondering if the diodes got hot enough to either fry or have extreme leakage current and slow switching to the point that it fried IGBTs without the diodes themselves failing.  Skip-pulse mode puts more heat into the diodes, as does secondary arcs.

Yes, DRSSTCs usually run at the lower of the dual resonance peaks of the coupled resonators, so a shorted secondary will raise the frequency.  The frequency change generally isn't significant enough to cause problems, but might be if your phase-shift circuit is particularly sensitive to frequency.  Usually the more significant effect is the increased Q of the primary, since the secondary no longer uses much power.  That causes the primary current to ramp up relatively quickly and ramp down slowly.  The pulse-skip mode will then need to skip lots of pulses, moving more power to the diodes, and likely causing more issues with gate-drive low-frequency components.  Even without pulse-skip, there will be more power for the diodes during the decay after the enable pulse ends.  (I'm working on a circuit to detect ground arcs and terminate the enable pulse before current ramps up too much.  My IGBTs have internal diodes, but their power dissipation capability is much lower than the IGBTs themselves.)

The other possible issue with external diodes is inductive voltage drop between the IGBTs and diodes.  I don't have any personal experience with external IGBT diodes.  I think IGBTs are usually rated for ~20V reverse Vce.  I don't know how sensitive they may be to spikes beyond 20V.

The "fuzzy" sine wave was likely multiple captures as the primary current ramps up.  The key reason to look there is for accurate phase adjustment.  Typically you can see the sine wave along with small steps at the top and bottom when the H-Bridge switches.  Those steps should be just barely before the top and bottom crests of the sine wave.  Try triggering on the enable pulse to get a stable waveform, or set the scope's trigger hold-off longer to prevent multiple triggers per enable pulse.

20
Yes, the fine phasing isn't that critical for SSTCs using FETs for the bridge.  It is critical for high-current DRSSTCs using IGBTs.  Still, efficiency is better with good phasing, and bridge layout (low inductance) is less critical with good phasing.

The gate-drive waveforms do look a lot like scope probe mis-adjustment.  However, the time scale is wrong for scope probe compensation.  I think your traces are likely accurate plots of UCC chip outputs.  They change rapidly until the voltage where they can't supply any more current to the gate series resistors.  Then they finish slewing as the gate capacitance is charged.

The bus supply should draw some current even if not sync'ed given the self-oscillation resistor addition - more than 18mA.  When sync'ed, it will increase several times.  At 15Vbus, the self-oscillation frequency will likely need to be adjusted closer to resonance (closer to 240kHz) to get sync'ing.  It should be possible to sync at 15V with good self-oscillation frequency.

The only reason I can imagine for FET frying at 30V is that there was some spike in gate-drive that over-voltaged the gates.  It's easy to get 24V from that gate-drive circuit, but that's generally not enough to fry FETs.  It would require a series resonance of the 0.1uF and gate-drive transformer inductance to go over 24V.  Perhaps somehow the antenna picked up the gate voltage and resonated at that frequency.  You could add a bidirectional TVS diode or back-to-back zener diodes from gate-to-emitter on each FET.  That would protect them from such gate-drive resonances.

Beyond adding FET gate voltage clamps, I don't know what other option you have other than starting to power it up.  I'd start with the current limit set low, perhaps 0.5 to 1A, just to be extra cautious.

Good luck!

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March 30, 2020, 02:10:26 AM

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